Electro-optical device, shift register circuit, and semiconductor device

ABSTRACT

An electro-optical device is configured to be capable of using a region of a gate line drive circuit efficiently and preventing rising speed of a gate line selection signal from decreasing (rising delay), and a shift register circuit is composed of a single conductivity type transistor which is suitable for the device. The gate line drive circuit including an odd driver to drive odd rows of a plurality of gate lines, and an even driver to drive even rows thereof. Each unit shift register in the odd and even drivers receives a selection signal in the second previous row and activates its own selection signal two horizontal periods later. A start pulse of the even driver is delayed in phase by one horizontal period with respect to a start pulse of the odd driver.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.12/705,235, filed Feb. 12, 2010, which is based upon and claims thebenefit of priority from prior Japanese Patent Application Nos.2009-027244, filed Feb. 9, 2009 and 2010-002580, filed Jan. 8, 2010.U.S. application Ser. No. 12/705,235 is incorporated herein byreference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to an electro-optical device such as animage display device and an image capturing apparatus and moreparticularly, to a scanning line drive circuit only composed of a sameconductivity type field-effect transistor, and a shift register circuitused therein.

2. Description of the Background Art

An electro-optical device having a scanning line drive circuit to scanpixels connected to a scanning line has been widely known. For example,an image display device (referred to as the “display device”hereinafter) such as a liquid crystal display device is configured suchthat a gate line (scanning an image display device (referred to as the“display device” hereinafter) such as a liquid crystal display device isconfigured such that a gate line (scanning line) is provided withrespect to each pixel row (pixel line) of a display element (displaypanel) in which a plurality of pixels are arranged in a form of amatrix, and a display image is updated by sequentially selecting anddriving the gate line every horizontal period of a display signal. Thegate line drive circuit (scanning line drive circuit) to sequentiallyselect and drive the pixel line, that is, the gate line employs a shiftregister to perform a shift operation which goes through a cycle for oneframe period of the display signal.

In addition, pixels of the imaging elements used in the image capturingapparatus are also arranged in the form of a matrix, and data of a takenimage is extracted when the gate line drive circuit scans those pixels.The shift register can be used in the gate line drive circuit of theimage capturing apparatus.

The shift register used in the gate line drive circuit is desirablycomposed of only the same conductivity type field-effect transistor toreduce the number of steps in a production process of the displaydevice. Therefore, various kinds of shift registers composed of only anN-type or P-type field-effect transistor and display devices having themare proposed (in the following Japanese Patent Application Laid-OpenNos. 2000-347628, 2004-78172, 2007-257813, and 2008-287753, forexample).

The shift register serving as the gate line drive circuit is composed ofa plurality of cascaded shift register circuits each provided in onepixel line, that is, one gate line. In this specification, each of theshift register circuits constituting the gate line drive circuit isreferred to as the “unit shift register”.

Japanese Patent Application Laid-Open No. 2000-347628 discloses a gateline drive circuit in which a stage (odd driver) to scan pixels in oddrows of multi-stage shift registers, and a stage (even driver) to scanpixels in even rows thereof are arranged so as to sandwich a displayelement or an imaging element. When the odd driver and the even driverare arranged in such a manner, a region for the gate line drive circuitcan be efficiently used.

FIGS. 3 and 4 of the above document show a unit shift register used inthe gate line drive circuit and a signal waveform thereof, respectively.In FIG. 3, a start signal IN inputted to a unit shift register RS1 o (1)in a first stage of an odd driver 2 o is temporally shifted by the unitshift register RS1 o (1) and outputted as a selection signal OUT1 of agate line GL1 of a first row of a liquid crystal display element. Theselection signal OUT1 is inputted to a unit shift register RS1 e (1) ina first stage of an even driver 2 e through the gate line GL1 andtemporally shifted by the unit shift register RS1 e (1), and outputtedas a selection signal OUT2 of a gate line GL2 of a second row. The sameoperation is performed in the unit shift registers in the followingstages, so that the gate lines are sequentially selected.

As shown in FIG. 3 of the above document, although the unit shiftregister in each stage has the same circuit configuration, here,attention is paid to the unit shift register RS1 e (1) in the firststage of the even driver 2 e. The selection signal OUT2 outputted fromthe unit shift register RS1 e (1) is activated when a clock signal/CK istransmitted to an output terminal by a transistor 204 which is turned onin response to activation of a selection signal OUT of the previousstage (unit shift register RS1 o(1)).

While the transistor 204 is turned on when wiring capacitances C2 and C4of its control electrode is charged by the selection signal OUT1, theselection signal OUT1 outputted from the previous stage is supplied tothe unit shift register RS1 e(1) through the gate line GL1, so that itis affected by a resistance component and a capacitance component of thegate line GL1. That is, rising speed of the selection signal OUT1decreases in proportion to a time constant based on the product of a sumof the resistance components and a sum of the capacitance components(this is described in paragraph 0043 in Japanese Patent ApplicationLaid-Open No. 2004-78172).

As a result, since charging speed of the control electrode of thetransistor 204 slows down, there is a concern that the control electrodeof the transistor 204 is not sufficiently charged when the shiftregister is operated at high speed. When the control electrode of thetransistor 204 is not sufficiently charged, on-resistance of thetransistor 204 increases, and charging speed of the output terminal,that is, rising speed of the selection signal OUT2 decreases.

The selection signal OUT2 is supplied to the next stage (unit shiftregister RS1 o (2) of the second stage in the odd driver 2 o) throughthe gate line GL2 and affected by a resistance component and acapacitance component of the gate line GL2 at this time, so that itsrising speed further decreases.

When this phenomenon is repeated every time the selection signal issequentially transmitted to the unit shift register in each stage, therising speed decreases as the selection signal goes through the latterstages, and the selection signal could not be activated in the middlestage and the selection signal could not be transmitted to the finalstage. This problem arises prominently in an electro-optical devicehaving a long gate line such as a display device having a large screen.

SUMMARY OF THE INVENTION

It is an object of the present invention to provide an electro-opticaldevice capable of using a region of a gate line drive circuitefficiently and preventing the rising speed of a gate line selectionsignal from decreasing (rising delay), and a shift register circuitcomposed of a single conductivity type transistor which is suitable forthe device.

An electro-optical device according to a first aspect of the presentinvention includes a scanning line drive circuit having an odd driverincluding a plurality of cascaded unit shift registers to drive oddlines of a plurality of scanning lines, and an even driver including aplurality of cascaded unit shift registers to drive even lines of theplurality of scanning lines. The odd and even drivers are formed on thesame substrate as the plurality of pixels and the plurality of scanninglines so as to sandwich the pixels. Each of the unit shift registers ofthe odd and even drivers has a first input terminal receiving an outputsignal of the unit shift register positioned in the second previous lineand operates so as to activate its own output signal delayed by ascanning period for two lines from an active period of the signalreceived by the first input terminal. A phase of the output signal ofthe odd driver is different from a phase of the output signal of theeven driver by a scanning period for one line.

An electro-optical device according to a second aspect of the presentinvention includes a scanning line drive circuit having an odd driverincluding a plurality of cascaded unit shift registers to drive oddlines of a plurality of scanning lines and capable of switching a shiftdirection of a signal, and an even driver including a plurality ofcascaded unit shift registers to drive even lines of the plurality ofscanning lines and capable of switching the shift direction of thesignal. The odd and even drivers are formed on the same substrate as theplurality of pixels and the plurality of scanning lines so as tosandwich the pixels. Each of the unit shift registers of the odd andeven drivers has a first input terminal receiving an output signal ofthe unit shift register positioned in the second previous line, and athird input terminal receiving an output signal of the unit shiftregister positioned in the second next line. Each shift registeroperates so as to activate its own output signal delayed by a scanningperiod for two lines from an active period of the signal received by thefirst input terminal at a time of forward shift, and operates so as toactivate its own output signal delayed by the scanning period for twolines from an active period of the signal received by the third inputterminal at a time of backward shift. A phase of the output signal ofthe odd driver is different from a phase of the output signal of theeven driver by a scanning period for one line.

According to the electro-optical device of the first and second aspectsof the present invention, the odd and even drives are arranged so as tosandwich pixels, whereby the region of the scanning line drive circuitcan be used efficiently. In addition, even in such a case, the outputsignal can be shifted without passing through a gate line. Therefore,the selection signal can be prevented from being delayed in rise becauseit is not affected by a resistance component and a capacitance componentof the gate line.

An electro-optical device according to a third aspect of the presentinvention includes a scanning line drive circuit having an odd driverincluding a plurality of cascaded unit shift registers to drive oddlines of the plurality of scanning lines, and an even driver including aplurality of cascaded unit shift registers to drive even lines of theplurality of scanning lines. Each of the unit shift registers of the oddand even drivers includes an output terminal, first and second inputterminals, a clock terminal, first and second transistors and a boostingunit, which are described below. The output terminal outputs an outputsignal of the unit shift register itself. The first input terminalreceives an output signal of the unit shift register positioned in thesecond previous line. The second input terminal receives a first clocksignal delayed in phase by a scanning period for one line with respectto the output signal of the unit shift register positioned in the secondprevious line. The clock terminal receives a second clock signal delayedin phase by the scanning period for one line from the first clocksignal. The first transistor supplies the second clock signal to theoutput terminal. The second transistor supplies the first clock signalto a first node connected to a control electrode of the firsttransistor. The boosting unit boosts a second node connected to acontrol electrode of the second transistor to a voltage higher than anamplitude of the first clock signal when the first clock signal isactivated following the activation of the signal received by the firstinput terminal. A phase of the output signal of the odd driver isdifferent from a phase of the output signal of the even driver by ascanning period for one line.

An electro-optical device according to a fourth aspect of the presentinvention includes a scanning line drive circuit having an odd driverincluding a plurality of cascaded unit shift registers to drive oddlines of the plurality of scanning lines and capable of switching ashift direction of a signal, and an even driver including a plurality ofcascaded unit shift registers to drive even lines of the plurality ofscanning lines and capable of switching the shift direction of thesignal. Each of the unit shift registers of the odd and even drivers hasan output terminal, first to fourth input terminals, a clock terminal,first to third transistors, and first and second boosting units, whichare described below. The output terminal outputs an output signal of theunit shift register itself. The first input terminal receives an outputsignal of the unit shift register positioned in the second previousline. The third input terminal receives an output signal of the unitshift register positioned in the second next line. The second inputterminal receives a first clock signal having the same phase as that ofan output signal of the unit shift register positioned in the previousline. The fourth input terminal receives a second clock signal havingthe same phase as that of an output signal of the unit shift registerpositioned in the next line. The clock terminal receives a third clocksignal delayed in phase by the scanning period for one line from thefirst clock signal at the time of forward shift and delayed in phase bythe scanning period for one line from the second clock signal at thetime of backward shift. The first transistor supplies the third clocksignal to the output terminal. The second transistor supplies the firstclock signal to a first node connected to a control electrode of thefirst transistor. The third transistor supplies the second clock signalto the first node. The first boosting unit boosts a second nodeconnected to a control electrode of the second transistor to a voltagehigher than an amplitude of the first clock signal when the first clocksignal is activated following the activation of the signal received bythe first input terminal at the time of forward shift. The secondboosting unit boosts a third node connected to a control electrode ofthe third transistor to a voltage higher than an amplitude of the secondclock signal when the second clock signal is activated following theactivation of the signal received by the third input terminal at thetime of backward shift. At the time of forward shift, the thirdtransistor of each of the unit shift registers of the odd and evendrivers is kept in an off state. At the time of backward shift, thesecond transistor of each of the unit shift registers of the odd andeven drivers is kept in an off state. A phase of the output signal ofthe odd driver is different from a phase of the output signal of theeven driver by a scanning period for one line.

According to the electro-optical device of the third and fourth aspectsof the present invention, in each shift register circuit, the boostingunit allows the transistor charging the first node to operate in theunsaturated region. Accordingly, the voltage between the gate and thesource of the first transistor can be further increased at high speed.Therefore, even when the frequency of the first clock signal is high,the driving capability of the first transistor, that is, the drivingcapability of the shift register circuit can be kept high, whichcontributes to higher speed operation. In addition, even when the oddand even drives are arranged so as to sandwich pixels to use the regionof the scanning line drive circuit efficiently, the output signal can beshifted without passing through a gate line. Therefore, the selectionsignal can be prevented from being delayed in rise because it is notaffected by a resistance component and a capacitance component of thegate line.

A shift register circuit according to a fifth aspect of the presentinvention includes first to fourth input terminals, an output terminal,and a clock terminal, first and second voltage signal terminals suppliedwith mutually complementary first and second voltage signals,respectively, first to seventh transistors, and first and second MOScapacitor elements. The first transistor supplies a clock signalinputted to the clock terminal to the output terminal. The secondtransistor supplies the first voltage signal to a first node connectedto a control electrode of the first transistor. The third transistorsupplies the second voltage signal to the first node. The fourthtransistor has a control electrode connected to the first input terminaland supplies the first voltage signal to a second node connected to acontrol electrode of the second transistor. The first MOS capacitorelement is connected between the second input terminal and the secondnode. The fifth transistor has a control electrode connected to thethird input terminal and supplies the second voltage signal to a thirdnode connected to a control electrode of the third transistor. Thesecond MOS capacitor element is connected between the fifth transistor,and the fourth input terminal and the third node. The sixth transistorhas a control electrode connected to the first input terminal andsupplies the first voltage signal to the first node. The seventhtransistor has a control electrode connected to the third input terminaland supplies the second voltage signal to the first node.

According to the shift register circuit of the present invention, sincethe second or third transistor charging the gate of the first transistoroperates in the unsaturated region, the voltage between the gate and thesource of the first transistor can be further increased at high speed.Therefore, even when the frequency of the first clock signal is high,the driving capability of the first transistor, that is, the drivingcapability of the shift register circuit can be kept high, whichcontributes to higher speed operation.

A semiconductor device according to a sixth aspect of the presentinvention includes a MOS capacitor element composed of an a-Si(amorphous silicon) transistor. The a-Si transistor serving as the MOScapacitor element has a gate functioning as one terminal of the MOScapacitor element, and at least one current electrode functioning as theother terminal of the MOS capacitor element. The gate length of the a-Sitransistor is longer than the gate width thereof.

A semiconductor device according to a seventh aspect of the presentinvention includes a MOS capacitor element composed of an a-Si(amorphous silicon) transistor. The a-Si transistor serving as the MOScapacitor element has a gate functioning as one terminal of the MOScapacitor element, and two current electrodes having different widths.Only the current electrode having a longer width of the two currentelectrodes is used as a terminal of the MOS capacitor element.

According to the semiconductor device of the present invention, the a-Sitransistor serving as the MOS capacitor element can be high in gatecapacitance functioning as the capacitor element and small in channelresistance while keeping its overlap capacitance between the gate andthe drain low. Therefore, the power consumption of the MOS capacitorelement can be cut and the response speed thereof can be increased.

These and other objects, features, aspects and advantages of the presentinvention will become more apparent from the following detaileddescription of the present invention when taken in conjunction with theaccompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic block diagram showing a configuration of a displaydevice according to the present invention;

FIG. 2 is a block diagram showing a configuration of a gate line drivecircuit according to a first embodiment;

FIG. 3 is a circuit diagram of a unit shift register according to thefirst embodiment;

FIG. 4 is a timing chart to describe an operation of the unit shiftregister according to the first embodiment;

FIG. 5 is a circuit diagram of a dummy unit shift register (first dummystage) in an odd driver according to the first embodiment;

FIG. 6 is a circuit diagram of a dummy unit shift register (second dummystage) in an even driver according to the first embodiment;

FIG. 7 is a timing chart showing an operation when the gate line drivecircuit according to the first embodiment is controlled with an endpulse;

FIG. 8 is a circuit diagram of a unit shift register according to asecond variation of the first embodiment;

FIG. 9 is a circuit diagram of a unit shift register according to athird variation of the first embodiment;

FIG. 10 is a circuit diagram of a unit shift register according to afourth variation of the first embodiment;

FIG. 11 is a circuit diagram of a unit shift register according to afifth variation of the first embodiment;

FIG. 12 is a block diagram showing a configuration of a gate line drivecircuit according to a second embodiment;

FIG. 13 is a circuit diagram of a unit shift register according to thesecond embodiment;

FIG. 14 is a circuit diagram of a unit shift register according to athird embodiment;

FIG. 15 is a circuit diagram of a unit shift register according to avariation of the third embodiment;

FIGS. 16A, 16B, and 16C are drawings showing a structure of a transistorserving as a MOS capacitor element according to a fourth embodiment;

FIG. 17 is a circuit diagram showing an example of a unit shift registerusing a MOS transistor according to the fourth embodiment;

FIG. 18 is a block diagram showing a configuration of a gate line drivecircuit according to a fifth embodiment;

FIG. 19 is a circuit diagram of a unit shift register according to asixth embodiment;

FIG. 20 is a timing chart to describe an operation of the unit shiftregister according to the sixth embodiment;

FIG. 21 is a circuit diagram of a unit shift register according to aseventh embodiment;

FIG. 22 is a block diagram showing a configuration of a gate line drivecircuit according to an eighth embodiment; and

FIG. 23 is another block diagram showing the configuration of the gateline drive circuit according to the eighth embodiment.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Embodiments of the present invention will be described hereinafter withreference to the drawings. In addition, to avoid the description frombeing reduplicated and redundant, the same reference is allocated to acomponent having the same or corresponding function in the drawings.

In addition, a transistor used in each embodiment is an insulated gatefield-effect transistor. In the insulated gate field-effect transistor,electric conductivity between a drain region and a source region in asemiconductor layer is controlled by an electric field in a gateinsulating film. The semiconductor layer in which the drain region andthe source region are formed may be made of an organic semiconductorsuch as polysilicon, amorphous silicon, and pentacene, and an oxidesemiconductor such as single-crystal silicon or IGZO (In—Ga—Zn—O).

As is well known, a transistor has at least three electrodes including acontrol electrode (gate (electrode) in a more limited sense), onecurrent electrode (drain (electrode) or source (electrode) in a morelimited sense), and the other current electrode (source (electrode) ordrain (electrode) in a more limited sense). The transistor functions asa switching element in which a channel is formed between the drain andsource when a predetermined voltage is applied to the gate. The drainand the source of the transistor have the same structure basically, andtheir names are switched depending on an applied voltage condition. Forexample, in a case of an N-type transistor, an electrode having arelatively high potential (also referred to as the “level” hereinafter)is called the drain, and an electrode having a relatively low potentialis called the source (vice versa in a case of a P-type transistor).

Those transistors may be formed on a semiconductor substrate, and may bea thin film transistor (TFT) formed on an insulating substrate such asglass unless otherwise stated. The substrate on which the transistor isformed may be a single-crystal substrate or an insulating substrate suchas SOI, glass, and a resin.

A gate line drive circuit according to the present invention only uses asingle conductivity type transistor. For example, the N-type transistorbecomes the active state (on state, or conductive state) when thevoltage between the gate and the source becomes the H (high) level whichis higher than the threshold voltage of the transistor, and becomes theinactive state (off state, or non-conductive state) when the voltagebetween them becomes the L (low) level which is lower than the abovethreshold voltage. Therefore, in a circuit using the N-type transistor,the H level of a signal is the “active level” and the L level thereof isthe “inactive level”. In addition, each node of the circuit using theN-type transistor changes from the inactive level to the active levelwhen charged to the H level, and changes from the active level to theinactive level when discharged to the L level.

On the other hand, the P-type transistor becomes the active state (onstate, or conductive state) when the voltage between the gate and thesource becomes the L level which is lower than the threshold voltage (anegative value based on the source) of the transistor, and becomes theinactive state (off state, or non-conductive state) at the high levelwhich is higher than the above threshold voltage. Therefore, in acircuit using the P-type transistor, the L level of a signal is the“active level” and the H level thereof is the “inactive level”. Inaddition, each node of the circuit using the P-type transistor changesfrom the inactive level to the active level when charged to the L level,and changes from the active level to the inactive level when dischargedto the H level, which is opposite in relationship between charge anddischarge to the case of the N-type transistor.

In this specification, the change from the inactive level to the activelevel and the change from the active level to the inactive level aredefined as “pull-up” and “pull-down”, respectively. In other words, thechange from the L level to the H level and the change from the H levelto the L level in the circuit using the N-type transistor are defined as“pull-up” and “pull-down”, respectively, and the change from the H levelto the L level and the change from the L level to the H level in thecircuit using the P-type transistor are defined as “pull-up” and“pull-down”, respectively.

Furthermore, in this specification, a description is made assuming thatthe “connection” between two elements, two nodes, or one element and onenode includes a state which is equivalent to substantially directconnection even when the connection is provided through another element(element or switch). For example, even when the two elements areconnected through the switch, the two elements are described as being“connected” as long as they can function similarly to the case wherethey are directly connected.

In the present invention, clock signals (multiphase clock signals)having different phases are used. Hereinafter, to simplify thedescription, a predetermined interval is provided between the activeperiod of one clock signal and the active period of another signal to beactivated next (interval between times t₂ and t₃ in FIG. 4, forexample). However, the above interval may not be provided as long as theactive period of each clock signal does not overlap substantially in thepresent invention. For example, when the active level is the H level,the falling timing of one clock signal may coincide with the risingtiming of the next clock signal.

First Embodiment

FIG. 1 is a schematic block diagram showing a configuration of a displaydevice according to the present invention, and shows an entireconfiguration of a liquid crystal display device as a representativeexample of the display device. In addition, the present invention is notlimited to the liquid crystal display device, and it may be widelyapplied to display devices to convert an electric signal to brightnessof light, such as an electroluminescence (EL), organic EL, plasmadisplay, and electronic paper, or electro-optical devices to convertintensity of light to an electric signal, such as an image capturingapparatus (imaging sensor).

A liquid crystal display device 100 includes a liquid crystal arraysection 10, a gate line drive circuit 30, and a source driver 40. Theliquid crystal array section 10 includes a plurality of pixels 15arranged in the form a matrix. Gate lines GL₁, GL₂ . . . (referred to asthe “gate line GL” collectively) are provided for rows of the pixels(also referred to as the “pixel line” hereinafter), respectively, anddata lines DL₁, DL₂, . . . (referred to as the “data line DL”collectively) are provided for columns of the pixels (also referred toas the “pixel column” hereinafter), respectively. More specifically, thepixel 15 is formed in the vicinity of the cross point between the gateline GL and the data line DL which intersect at right angles to eachother. FIG. 1 shows the pixels 15 in the first and second columns alongthe first row, and corresponding gate lines GL₁ and GL₂, and the datalines DL₁ and DL₂, representatively.

Each pixel 15 has a pixel switch element 16 provided between thecorresponding data line DL and a pixel node Np, and a capacitor 17 and aliquid crystal display element 18 connected in parallel between thepixel node Np and a common electrode node Nc. The orientation of theliquid crystal in the liquid crystal display element 18 changes based ona voltage difference between the pixel node Np and the common electrodenode Nc, and accordingly, the display brightness of the liquid crystaldisplay 18 changes. Thus, the brightness of the pixel can be controlledby a display voltage transmitted through the data line DL and the pixelswitch element 16. That is, intermediate brightness can be obtained byapplying an intermediate voltage difference between a voltage differencecorresponding to a maximum brightness and a voltage differencecorresponding to a minimum brightness, to between the pixel node Np andthe common electrode node Nc. Therefore, gradational brightness can beobtained by setting the above display voltage in stages.

The gate line drive circuit 30 is composed of an odd gate line drivecircuit (odd driver) 30 a to drive the gate lines GL₁, GL₃, GL₅ . . . inthe odd rows, and an even gate line drive circuit (even driver) 30 b todrive the gate lines GL₂, GL₄, GL₆ . . . in the even rows, andsequentially selects and activates the gate lines GL, based on apredetermined scanning cycle. The gate electrode of the pixel switchelement 16 is connected to the corresponding gate line GL. While thespecific gate line GL is selected, the pixel switch element 16 is in theconductive state in the pixel connected thereto and the pixel node Np isconnected to the corresponding data line DL. Thus, the display voltagetransmitted to the pixel node Np is kept in the capacitor 17. Ingeneral, the pixel switch element 16 is composed of a TFT formed on thesame insulating substrate (glass substrate, resin substrate, and thelike) as the liquid crystal display element 18.

The source driver 40 outputs the display voltage set in stages by adisplay signal SIG serving as an N-bit digital signal to the data lineDL. Here, as one example, it is assumed that the display signal SIG is a6-bit signal composed of display signal bits DB0 to DB5. Based on the6-bit display signal SIG, gradation display in 2⁶=64 stages can beprovided in each pixel. Furthermore, when one color display unit isformed of three pixels of R (Red), G (Green), and B (Blue), about260,000 colors can be displayed.

In addition, as shown in FIG. 1, the source driver 40 is composed of ashift register 50, data latch circuits 52 and 54, a gradational voltagegeneration circuit 60, a decode circuit 70, and an analog amplifier 80.

The display signal bits DB0 to DB5 corresponding to the displaybrightness of each pixel are serially generated in the display signalSIG. That is, the display signal bits DB0 to DB5 at each timing show thedisplay brightness in one pixel 15 in the liquid crystal array section10.

The shift register 50 orders the data latch circuit 52 to load thedisplay signal bits DB0 to DB5 at a timing synchronized with a cyclewhen the setting of the display signal SIG is switched. The data latchcircuit 52 sequentially loads the serially generated display signal SIGand stores the display signal SIG for the one pixel line.

A latch signal LT inputted to the data latch circuit 54 is activatedwhen the display signal SIG for the one pixel line is loaded by the datalatch circuit 52. In response to this, the data latch circuit 54 loadsthe display signal SIG for the one pixel line stored in the data latchcircuit 52.

The gradational voltage generation circuit 60 is composed of 63 voltagedividing resistors connected between a high voltage VDH and a lowvoltage VDL in series, and generates 64-stage gradational voltages V1 toV64.

The decode circuit 70 decodes the display signal SIG stored in the datalatch circuit 54, and selects the voltage from the gradational voltagesV1 to V64 based on the above decoded result and outputs it to the decodeoutput nodes Nd₁, Nd₂ . . . (referred to as the “decode output node Nd”correctively).

As a result, the display voltage (one of the gradational voltages V1 toV64) corresponding to the display signal SIG for the one pixel linestored in the data latch circuit 54 is outputted to the decode outputnode Nd at the same time (in parallel). In addition, in FIG. 1, thedecode output nodes Nd₁ and Nd₂ corresponding to the data lines DL₁ andDL₂ in the first and second columns, respectively are representativelyshown.

The analog amplifier 80 amplifies a current of an analog voltagecorresponding to the display voltage outputted from the decode circuit70 to each of the decode output nodes Nd₁, Nd₂ . . . and outputs it toeach of the data lines DL₁, DL₂ . . . .

The source driver 40 repeatedly outputs the display voltagecorresponding to the series of display signal SIG to the data line DLwith respect to each pixel line, based on a predetermined scanningcycle, and the gate line drive circuit 30 sequentially drives the gatelines GL₁, GL₂ . . . in synchronization with the scanning cycle, wherebyan image is displayed in the liquid crystal array section 10 based onthe display signal SIG.

In addition, while FIG. 1 illustrates the configuration of the liquidcrystal display device 100 in which the gate line drive circuit 30 andthe source driver 40 are integrated with the liquid crystal arraysection 10, as another configuration, the gate line drive circuit 30 andthe liquid crystal array section 10 may be formed integrally and thesource driver 40 may be provided as an external circuit outside theliquid crystal array section 10, or both of the gate line drive circuit30 and the source driver 40 may be provided as external circuits outsidethe liquid crystal array section 10.

FIG. 2 is a drawing showing a configuration of the gate line drivecircuit 30 according to the present invention. The gate line drivecircuit 30 has unit shift registers SR₁, SR₂, SR₃ . . . (the unit shiftregisters SR₁, SR₂, SR₃ . . . are also referred to as the “unit shiftregister SR” collectively) provided with respect to each pixel line,that is, each gate line GL.

As described above, the gate line drive circuit 30 includes the odddriver 30 a and the even driver 30 b. The odd driver 30 a includes thecascaded unit shift registers SR₁, SR₃, SR₅ . . . to drive the gatelines GL₁, GL₃, GL₅ . . . in the odd rows. The even driver 30 b includesthe cascaded unit shift registers SR₂, SR₄, SR₆ . . . to drive the gatelines GL₂, GL₄, GL₆ . . . in the even rows. More specifically, in eachof the odd driver 30 a and the even driver 30 b, when viewed from akth-stage unit shift register SR_(k), a “previous stage” means a unitshift register SR_(k−2) in the second previous row, and a “next stage”means a unit shift register SR_(k+2) in the second next row.

The odd driver 30 a according to this embodiment is provided with adummy unit shift register SRD1 (referred to as the “first dummy stage”hereinafter) connected to a dummy gate line DML which does not drive thepixel, at the next stage of the last-stage unit shift register SR_(n-1).On the other hand, the even driver 30 b according to this embodiment isprovided with a dummy unit shift register SRD2 (referred to as the“second dummy stage” hereinafter) which is not connected to the gateline, at the next stage of the last-stage unit shift register SR_(n).The first and second dummy stages SRD1 and SRD2 have the same circuitconfiguration as the other unit shift register SR basically, and theirdetailed configurations will be described below.

In addition, a clock signal generator 31 shown in FIG. 2 supplies threeclock signals CLK1, CLK2, and CLK3 having different phases from eachother, to the unit shift registers SR in the gate line drive circuit 30.The active periods of the clock signals CLK1 to CLK3 do not overlap witheach other and the CLK1, CLK2, CLK3, CLK1, CLK2 . . . are controlled soas to be repeatedly activated in this order at the timing synchronizedwith the scanning cycle of the display device.

Each unit shift register SR has a first input terminal IN1, a secondinput terminal IN2, an output terminal OUT, a clock terminal CK, and areset terminal RST. As shown in FIG. 2, the clock terminal CK of eachunit shift register SR is supplied with a predetermined one of the clocksignals CLK1, CLK2, and CLK3 outputted from the clock signal generator31.

More specifically, the clock signal CLK1 is supplied to the unit shiftregisters SR₁, SR₄, SR₇ . . . which drive the gate lines GL_(3m-2) inthe [3m−2]th rows (m is a natural number and the same is appliedhereinafter). The clock signal CLK2 is supplied to the unit shiftregisters SR₂, SR₅, SR₈ . . . which drive the gate lines GL_(3m-1) inthe [3m−1]th rows. The clock signal CLK3 is supplied to the unit shiftregisters SR₃, SR₆, SR₉ . . . which drive the gate lines GL_(3m) in the[3m]th rows. Since the clock signals CLK1, CLK2, and CLK3 are activatedin this order, the clock terminals CK of the shift registers SR₁, SR₂,SR₃ . . . are activated in this order.

In addition, since the number of scanning lines of the general displaydevice is not multiples of three, in the shift register controlled bythe three clock signals CLK1 to CLK3, the clock signal supplied to theclock terminal CK of the last unit shift register SR_(n) in the last nthrow (the last stage of the even driver 30 b) differs depending on thenumber of the scanning lines of the display device. In FIG. 2, the clocksignal CLK2 is supplied to the clock terminal CK of the unit shiftregister SR_(n). Therefore, the clock signals CLK3 and CLK1 are suppliedto the clock terminals CK of the first and second dummy stages SRD1 andSRD2, respectively.

First and second start pulses SP1 and SP2 are inputted to the first andsecond input terminals IN1 and IN2 of the unit shift register SR₁ in thefirst row (first stage of the odd driver 30 a), respectively. Accordingto this embodiment, while both of the first and second start pulses SP1and SP2 are activated (become the H level) at the timing correspondingto the head of each frame period of the image signal, the second startpulse SP2 is delayed in phase with respect to the first start pulse SP1by one horizontal period (1H), that is, by the scanning period for theone line.

Therefore, the first start pulse SP1 becomes the H level earlier thanthe second start pulse SP2, and the second start pulse SP2 is controlledso as to become the H level after the first start pulse SP1 has returnedto the L level. Here, it is assumed that the first start pulse SP1 hasthe same phase as the clock signal CLK2, and the second pulse SP2 hasthe same phase as the clock signal CLK3 (refer to FIG. 4).

In addition, the second start pulse SP2 is inputted to the first inputterminal IN1 of the unit shift register SR₂ in the second row (firststage of the even driver 30 b). Then, the clock signal CLK1 activatedafter one horizontal period has elapsed from the second start pulse SP2is inputted to the second input terminal IN2 thereof.

In the unit shift register SR_(k) after the third row, the first inputterminal IN1 is connected to the output terminal OUT of the unit shiftregister SR_(k−2) positioned in the second previous row thereof (theprevious stage in the odd driver 30 a or the even driver 30 b). Inaddition, the second input terminal IN2 is supplied with the clocksignal delayed in phase by one horizontal period with respect to thesignal inputted to the first input terminal IN1 (outputted from the unitshift register SR_(k−2)) (this is advanced in phase by one horizontalperiod with respect to the one supplied to the clock signal CK).

Similarly, in the first and second dummy stages SRD1 and SRD2, the firstinput terminal IN1 of the first dummy stage SRD1 is supplied with anoutput signal G_(n-1) of the unit shift register SR_(n-1) positioned inthe second previous row, and the second input terminal IN2 thereof issupplied with the clock signal CLK2 delayed in phase by one horizontalperiod with respect to the output signal G_(n-1). In addition, the firstinput terminal IN1 of the second dummy stage SRD2 is supplied with anoutput signal G_(n) of the unit shift register SR_(n) positioned in thesecond previous row, and the second input terminal IN2 thereof issupplied with the clock signal CLK3 delayed in phase by one horizontalperiod with respect to the output signal G_(n).

The reset terminal RST of the unit shift register SR_(k) is connected tothe output terminal OUT of the unit shift register SR_(k+2) positionedin the second next row (the next row in the odd driver 30 a or the evendriver 30 b) (the reset terminal RST may be connected to the outputterminal OUT of the unit shift register SR_(k+1) positioned in the nextrow, but it is preferable that it is connected to the output terminalOUT of the unit shift register SR_(k+2) positioned in the second nextrow. This reason will be described below).

The reset terminal RST of the unit shift register SR_(n-1) in thenext-to-last row (last stage of the odd driver 30 a) is connected to theoutput terminal OUT of the first dummy stage SRD1 positioned in thesecond next row. In addition, the reset terminal RST of the last unitshift register SR_(n) (last stage of the even driver 30 b) is connectedto the output terminal OUT of the second dummy stage SRD2.

Furthermore, the reset terminal RST of the first dummy stage SRD1 issupplied with the clock signal CLK2 delayed in phase by two horizontalperiods with respect to the clock signal CLK3 supplied to the clockterminal CK. The reset terminal RST of the second dummy stage SRD2 issupplied with the clock signal CLK3 delayed in phase by two horizontalperiods with respect to the clock signal CLK1 supplied to the clockterminal CK.

As described above, the output signal G_(k) outputted from the outputterminal OUT of the unit shift register SR_(k) is supplied to thecorresponding gate line GL_(k) as a selection signal (vertical (orhorizontal) scanning pulse), and also supplied to the first inputterminal IN1 of the unit shift register SR_(k+2) in the second next rowand to the reset terminal RST of the unit shift register SR_(k−2) in thesecond previous row. Hereinafter, the output signal G of the unit shiftregister SR is referred to as the “selection signal”.

FIG. 3 is a circuit diagram showing a configuration of the unit shiftregister according to the first embodiment of the present invention.Since the configuration of the unit shift register SR of the gate linedrive circuit 30 is substantially the same, the unit shift registerSR_(k) in the kth row is representatively shown. As all transistors ofthe unit shift register SR_(k) according to this embodiment are the sameconductivity type field-effect transistors, it is assumed that theN-type TFT is used in the following embodiments and variation.

The unit shift register SR_(k) shown in FIG. 3 is provided by applyingthe circuit in FIG. 6 in Japanese Patent Application Laid-Open No.2007-257813 according to the idea of the inventor of this application,to the present invention. The unit shift register SR_(k) has a firstpower supply terminal S1 supplied with a low potential side power supplypotential (low side power supply potential) VSS and second and thirdpower supply terminals S2 and S3 supplied with high potential side powersupply potentials (high side power supply potentials) VDD1 and VDD2,respectively, as well as the first and second input terminals IN1 andIN2, the output terminal OUT, the clock terminal CK and the resetterminal RST shown in FIG. 2.

The high side power supply potentials VDD1 and VDD2 may be at the samelevel. The description will be made assuming that the low side powersupply potential VSS is set to a reference potential (VSS=0V) of thecircuit, but in practice, the reference potential is set based on thevoltage of data to be written in the pixel, and the high side powersupply potentials VDD1 and VDD2 are set to 17 V and the low side powersupply potential VSS is set to −12 V, for example.

The unit shift register SR_(k) is composed of an output circuit 20, apull-up drive circuit 21, and a pull-down drive circuit 22. The outputcircuit 20 activates and inactivates the selection signal G_(k)outputted from the unit shift register SR_(k), and includes a transistorQ1 (output pull-up transistor) which puts the selection signal G_(k)into the active state (H level) while the gate line GL_(k) is selectedand a transistor Q2 (output pull-down transistor) which keeps theselection signal G_(k) in the inactive state (L level) while the gateline GL_(k) is not selected.

The transistor Q1 is connected between the output terminal OUT and theclock terminal CK, and activates the selection signal G_(k) by supplyingthe clock signal inputted to the clock terminal CK to the outputterminal OUT. In addition, the transistor Q2 is connected between theoutput terminal OUT and the first power supply terminal S1, and keepsthe selection signal G_(k) at the inactive level by discharging theoutput terminal OUT to the potential VSS. Here, a node connected to thegate (control electrode) of the transistor Q1 is defined as a “node N1”and a node connected to the gate of the transistor Q2 is defined as a“node N2”.

A capacitor element C1 is provided between the gate and the source ofthe transistor Q1 (that is, between the output terminal OUT and the nodeN1). This capacitor element C1 capacitively couples the output terminalOUT with the node N1 to enhance a boost effect of the node N1 associatedwith the elevation in level of the output terminal OUT. Here, it is tobe noted that the capacitor element C1 can be replaced with thetransistor Q1 when the capacitance between the gate and the channel issufficiently high, so that it may be omitted in that case.

In general, since a thickness of an insulating film serving as adielectric layer of a capacitor element is the same as that of a gateinsulating film of a transistor in one semiconductor integrated circuit,the capacitor element can be replaced with the transistor having thesame gate area as that of the capacitor element when the capacitorelement is substituted with the gate capacitance of the transistor.Thus, when the capacitor element C1 shown in FIG. 3 is replaced with thecapacitance between the gate and the channel of the transistor Q1, thegate width of the transistor Q1 is to be increased by a correspondingamount.

The pull-up drive circuit 21 drives the transistor Q1 (output pull-uptransistor), and puts the transistor Q1 in the on state while the gateline GL_(k) is selected, and puts the transistor Q1 in the off statewhile it is not selected. The pull-up drive circuit 21 charges the nodeN1 (the gate of the transistor Q1) in response to the activation of theselection signal G_(k−2) inputted from the second previous row to thefirst input terminal IN1 (or the first or second start pulse SP1 or SP2)and the clock signal (CLK1, CLK2, or CLK3) inputted to the second inputterminal IN2, and discharges the node N1 in response to the activationof the selection signal G_(k+2) in the second next row, serving as thereset signal supplied to the reset terminal RST (or the output signal D1or D2 of the first dummy stage SRD1 or the second dummy stage SRD2).

The pull-up drive circuit 21 is composed of transistors Q3 to Q5 and Q8to Q10. The transistor Q3 is connected between the node N1 and thesecond power supply terminal S2, and supplies the potential of thesecond power supply terminal S2 to the node N1. Here, a node connectedto the gate of the transistor Q3 is defined as a “node N3”.

The transistor Q4 is connected between the node N1 and the first powersupply terminal S1, and its gate is connected to the node N2. Thetransistor Q8 is connected between the node N3 and the second powersupply terminal S2, and its gate is connected to the first inputterminal IN1. In addition, the drain of the transistor Q8 may beconnected to the first input terminal IN1 (that is, the transistor Q8may be diode-connected between the first input terminal IN1 and the nodeN3).

The transistor Q10 has a gate connected to the node N3 and two currentelectrodes (source and drain) connected to the second input terminalIN2. The field-effect transistor is an element which is turned on whenthe drain and source thereof are electrically connected through theconductive channel formed just under the gate electrode through the gateinsulating film in the semiconductor substrate when a voltage higherthan the threshold voltage is applied to the gate electrode. Therefore,the on-state field-effect transistor has certain electrostaticcapacitance (gate capacitance) between the gate and the channel. Thatis, it can function as a capacitor element in which the channel in thesemiconductor substrate and the gate electrode serve as both terminalsand the gate insulating film serves as a dielectric layer. Therefore,the transistor Q10 selectively serves as the capacitor element based onthe voltage between the node N3 and the second input terminal IN2(functions as the capacitor element only while the node N3 is at the Hlevel).

In addition, since the second start pulse SP2 supplied to the secondinput terminal IN2 of the unit shift register SR₁ in the first row isactivated only once for one frame period, it does not necessarily serveas the capacitor element selectively (or may always function as thecapacitor element). Therefore, a usual capacitor element may be usedinstead of the MOS capacitor element (transistor Q10) in the unit shiftregister SR₁.

The transistor Q5 is connected between the node N3 and the first powersupply terminal S1, and its gate is connected to the reset terminal RST.The transistor Q9 is connected between the node N3 and the first powersupply terminal S1, and its gate is connected to the node N2.

On the other hand, the pull-down drive circuit 22 drives the transistorQ2 (output pull-down transistor) and its input end is the node N3 andits output end is the node N2 (the gate of the transistor Q2). That is,the pull-down drive circuit 22 charges/discharges the node N2 inresponse to the level change of the node N3. More specifically, itdischarges the node N2 when the node N3 becomes the H level, and chargesthe node N2 when the node N3 becomes the L level. Thus, the transistorQ2 is in the off state while the gate line GL_(k) is selected and in theon state while it is not selected. In addition, as described above, thenode N2 serving as the output end of the pull-down drive circuit 22 isconnected to the gates of the transistors Q4 and Q9 in the pull-up drivecircuit 21.

The pull-down drive circuit 22 is composed of transistors Q6 and Q7connected in series between the third power supply terminal S3 and thefirst power supply terminal S1. The transistor Q6 is connected betweenthe node N2 and the third power supply terminal S3 and its gate isconnected to the third power supply terminal S3 (that is, the transistorQ6 is diode-connected). The transistor Q7 is connected between the nodeN2 and the first power supply terminal S1 and its gate is connected tothe node N3.

The transistor Q7 is set such that its on-resistance is sufficientlysmall (that is, driving capability is high) as compared with thetransistor Q6. Therefore, when the gate (node N3) of the transistor Q7becomes the H level and the transistor Q7 is turned on, the node N2 isdischarged to the L level, while when the node N3 becomes the L leveland the transistor Q7 is turned off, the node N2 becomes the H level.

That is, the pull-down drive circuit 22 serves as a ratio-type inverterwhose operation is defined by a ratio between the on-resistance valuesof the transistor Q6 and the transistor Q7. In this inverter, thetransistor Q6 functions as a load element, and the transistor Q7functions as a drive element.

A description will be made of a specific operation of the unit shiftregister SR according to this embodiment. Since the operations of eachof the unit shift registers SR and the dummy stage SRD in the gate linedrive circuit 30 are substantially the same, the operation of the unitshift register SR_(k) in the kth row will be described representatively.It is assumed that the clock signal CLK1 is inputted to the clockterminal CK in the unit shift register SR_(k) (for example, thiscorresponds to the unit shift registers SR₁, SR₄ . . . at 3m−2 stages inFIG. 2).

To simplify the description, it is assumed that potentials at the Hlevel of the clock signals CLK1 to CLK3, and the first and second startpulses SP1 and SP2 are all the same and the level is set to VDD unlessotherwise stated. In addition, the VDD is also equal to the levels ofthe high side power supply potentials VDD1 and VDD2 (that is,VDD=VDD1=VDD2). In addition, potentials at the L level of the clocksignals CLK1 to CLK3, and the first and second start pulses SP1 and SP2are equal to the low side power supply potential VSS and theirpotentials are set to 0V (VSS=0). Furthermore, it is assumed that thethreshold voltages of the transistors are all equal to each other, andits value is set to Vth. Still furthermore, the clock signals CLK1 toCLK3 are repetition signals having phase differences of one horizontalperiod (1H) as shown in FIG. 4).

FIG. 4 is a timing chart to describe the operation of the unit shiftregister according to the first embodiment. The operation of the unitshift register SR_(k) will be described with reference to the drawing.

It is assumed that as an initial state, the nodes N1 and N3 are at the Llevel (VSS), and the node N2 is at the H level (VDD−Vth) (this state isreferred to as the “reset state” hereinafter). In addition, it isassumed that the first input terminal IN1 (selection signal G_(k−2) inthe second previous row), the second input terminal IN2 (clock signalCLK3), the clock terminal CK (clock signal CLK1), and the reset terminalRST (selection signal G_(k+2) in the second next row) are all at the Llevel.

Since the transistor Q1 is in the off state (cut-off state) and thetransistor Q2 is in the on state (conductive state) in the reset state,the output terminal OUT (selection signal G_(k)) is kept at the L levelregardless of the level of the clock terminal CK (clock signal CLK1).That is, in this initial state, the gate line GL_(k) corresponding tothe unit shift register SR_(k) is in the unselected state.

When the selection signal G_(k−2) in the second previous row (the firststart pulse SP1 in the case of the unit shift register SR₁ in the firstrow) becomes the H level at a time t₁ from the above state, thetransistor Q8 of the unit shift register SR_(k) is turned on. At thistime, the transistor Q9 is also on because the node N2 is at the Hlevel, but since the on-resistance of the transistor Q8 is set so as tobe sufficiently lower than that of the transistor Q9, the node N3 ischarged with electric charges supplied through the transistor Q8 and itslevel rises. That is, the transistor Q8 functions as a charge circuit tocharge the node N3 connected to the gate of the transistor Q3, based onthe signal inputted to the first input terminal IN1.

When the level of the node N3 rises, the transistor Q7 becomesconductive and the level of the node N2 falls. This causes theresistance of the transistor Q9 to increase and the level of the node N3rises rapidly. Accordingly, the transistor Q7 is sufficiently turned on.As a result, the node N2 becomes the L level (VSS), and the transistorQ9 is turned off, and the node N3 becomes the H level.

It is necessary to charge the capacitances between the gates and thechannels (gate capacitances) of the transistor Q10 and transistor Q3 toraise the level of the node N3, but since their capacitance values areas small as about one-fifth to one-tenth of the gate capacitance of thetransistor Q1 of the output circuit 20 and the capacitor element C1, thenode N3 can be charged at high speed. Therefore, the level of the nodeN3 rises to a theoretical value at high speed even though the transistorQ8 operates in a source follower mode in which it is no good athigh-speed charging. That is, a potential V3 a of the node N3 aftercharged by the transistor Q8 is as follows.V3a=VDD−Vth  (1)

When the node N3 becomes the H level, the transistor Q3 is turned on inresponse to it. At this time, since the node N2 is at the L level, thetransistor Q4 is in the off state and the level of the node N1 rises.

It is necessary to charge the capacitor element C1 and the gatecapacitance of the transistor Q1 in order to raise the level of the nodeN1, but since their capacitance values are relatively great as describedabove, it is difficult to charge the node N1 at high speed. Furthermore,since the transistor Q3 operates in the source follower mode, it isdifficult to raise the level of the node N1 to a theoretical value(VDD−2×Vth) for a short time. Therefore, when a pulse width of theselection signal G_(k−2) in the second previous row is not sufficientlylarge, the level of the node N1 at that time only rises to a level lowerthan the theoretical value.

When the selection signal G_(k−2) in the second previous row returns tothe L level at a time t₂, the nodes N1 and N3 become a floating statewhile the transistor Q8 is turned off, and the transistors Q7 and Q9serve as flip-flop elements, so that the levels of the nodes N1 and N3are maintained.

Then, when the clock signal CLK3 (the second start pulse SP2 in the caseof the unit shift register SR₁ in the first row) becomes the H level ata time t₃, the second input terminal IN2 of the unit shift registerSR_(k) becomes the H level. At this time, since the node N3 is at the Hlevel, a channel is formed between the source and the drain (on the sideof IN2) of the transistor Q10. Therefore, the transistor Q10 serves asthe capacitor element, and the node N3 is boosted by the capacitivecoupling through it. That is, the transistor Q10 functions as a boostercircuit in which the charged node N3 is boosted, based on the signalinputted to the first input terminal IN1.

When it is assumed that the parasitic capacitance of the node N3 issufficiently lower than the capacitance value of the transistor Q10serving as the MOS capacitor element, the node N3 after boosted by thetransistor Q10 rises from the potential V3 a before boosted by theamplitude VDD of the clock signal CLK3. That is, the potential V3 b ofthe node N3 after boosted is as follows.V3b=2×VDD−Vth  (2)Additionally, since the node N3 is boosted according to the clock signalCLK3 which is an external signal having high rising speed, the risingspeed of the potential of the node N3 is as high as the rising speed ofthe clock signal CLK3.

When the node N3 is boosted, the voltage between the gate (node N3) andthe source (node N1) of the transistor Q3 becomes sufficiently high, sothat the transistor Q3 operates in an unsaturated region instead ofoperating in the source follower mode and charges the node N1. Thus, thenode N1 is charged at high speed and the node N1 reaches the potentialVDD without losing the threshold voltage (Vth) of the transistor Q3.Thus, when the nodes N1 and N3 become the H level, and the node N2becomes the L level (this state is referred to as the “set state”hereinafter), the transistor Q1 is turned on and the transistor Q2 isturned off.

When the clock signal CLK 3 returns to the L level at a time t₄, thepotential of the node N3 is lowered by the transistor Q10 serving as theMOS capacitor element, and returns to the VDD−Vth before boosted. Atthis time, since the node N1 is at the potential VDD, the node N1 is inthe floating state and maintained at the potential VDD while thetransistor Q3 is turned off. Therefore, the set state of the unit shiftregister SR_(k) is maintained.

When the clock signal CLK1 becomes the H level at a time t₅, thatpotential change is transmitted to the output terminal OUT through thetransistor Q1 in the on state, and the level of the selection signalG_(k) rises. At this time, the level of the node N1 is boosted by aspecific amount by capacitive coupling through the capacitor element C1and the gate capacitance of the transistor Q1 (therefore, the node N1 issometimes called a “booster node”).

Therefore, even while the level of the selection signal G_(k) is rising,the voltage between the gate and the source of the transistor Q1 is kepthigh, and the transistor Q1 operates in the unsaturated region. Thus,the output terminal OUT is charged at high speed, and the level of theselection signal G_(k) rises at high speed following the rise of theclock signal CLK1. As a result, the level of the selection signal G_(k)reaches the VDD similarly to the clock signal CLK1 without losing thethreshold voltage Vth of the transistor Q1.

In addition, when it is assumed that the parasitic capacitance value ofthe node N1 is sufficiently small as compared with the sum ofcapacitance values of the gate capacitance of the transistor Q1 and thecapacitor element C1, the boost width of the node N1 is VDD which is thesame as the amplitude of the clock signal CLK1 and the selection signalG_(k). Therefore, the potential of the node N1 after boosted is 2×VDD.

Then, while the clock signal CLK1 is at the H level (for times t₅ tot₆), the selection signal G_(k) is kept at the H level. Therefore,during that period, the gate line GL_(k) is activated and becomes theselected state.

Then, when the clock signal CLK1 returns to the L level at the time t₆,the output terminal OUT is discharged through the transistor Q1, and theselection signal G_(k) becomes the L level. Thus, the gate line GL_(k)is inactivated and returns to the unselected state. At this time, thelevel of the node N1 also returns to the VDD before boosted.

Then, while the clock signal CLK2 is at the H level for times t₇ to t₈,the clock signal CLK2 is not supplied to the unit shift register SR_(k),so that there is no level change of each node of that unit shiftregister SR_(k).

However, at this time, since the selection signal G_(k+1) outputted fromthe unit shift register SR_(k+1) positioned in the next row isactivated, the unit shift register SR_(k+2) positioned in the secondnext row moves to the set state. Thus, when the clock signal CLK3becomes the H level at a time t₉, the selection signal G_(k+2) in thesecond next row becomes the H level.

Thus, in the unit shift register SR_(k) at the time t₉, the resetterminal RST becomes the H level and the transistor Q5 is turned on.Then, the node N3 is discharged and becomes the L level and thetransistor Q7 is turned off, so that the node N2 becomes the H level.Accordingly, the transistors Q4 and Q9 are turned on and the nodes N1and N3 become the L level. That is, the unit shift register SR_(k)returns to the reset state and the transistor Q1 is turned off and thetransistor Q2 is turned on.

In addition, when the clock signal CLK3 rises at the time t₉, the levelof the node N3 tries to rise through the transistor Q10 serving as theMOS capacitor element, but since the transistor Q5 is turned on almostat the same time, the level rises just momentarily and the node N3 iskept roughly at the L level.

On and after the time t₉, the transistors Q7 and Q9 function asflip-flop elements and the node N2 is kept at the H level, and the nodeN3 is kept at the L level. In addition, since the transistor Q10 doesnot form the channel and does not function as the capacitor elementwhile the node N3 is at the L level, the node N3 is not boosted and keptat the L level even when the clock signal CLK3 of the second inputterminal IN2 is activated after that. Therefore, the unit shift registerSR_(k) is kept in the reset state until the selection signal G_(k−2) inthe second previous row is activated in the next frame period.

The above is summarized as follows. The unit shift register SR_(k) iskept in the reset state until the signal of the first input terminal IN1is activated, and the selection signal G_(k) is kept at the L level.Then, when the signal of the first input terminal IN1 becomes the Hlevel, the node N3 is charged and the transistor Q3 is turned on tocharge the node N1, so that the unit shift register SR_(k) moves to theset state. Then, when the signal of the second input terminal IN2becomes the H level, the node N3 is boosted and the transistor Q3operates in the unsaturated region, so that the potential of the node N1rises to VDD. Then, when the signal of the clock terminal CK becomes theH level, the output terminal OUT is charged by the transistor Q1 in theon state, and the selection signal G_(k) is activated. Then, when thesignal of the reset terminal RST becomes the H level, the unit shiftregister SR_(k) returns to the reset state and the selection signalG_(k) is kept at the L level again.

When the odd driver 30 a and the even driver 30 b are configured asshown in FIG. 2, in each unit shift register SR_(k), the first inputterminal IN1 is supplied with the selection signal G_(k−2) in the secondprevious row, the second input terminal IN2 is supplied with the clocksignal which is activated one horizontal period later than the selectionsignal G_(k−2) in the second previous row (the clock signal having thesame phase as the selection signal G_(k−1) in the previous row (the samesignal as that of the clock terminal CK of the unit shift registerSR_(k−1) in the previous row)), and the clock terminal CK is suppliedwith the clock signal which is activated two horizontal periods laterthan the selection signal G_(k−2) in the second previous row. Inaddition, the reset terminal RST is supplied with the selection signalG_(k+2) in the second next row.

Thus, each unit shift register SR_(k) in the odd driver 30 a and theeven driver 30 b becomes the set state in response to the activation ofthe selection signal G_(k−2) in the second previous row, and thepotential of the node N1 is raised to VDD after one horizontal period,and the selection signal G_(k) is activated after two horizontalperiods. That is, each unit shift register SR_(k) operates to activateits selection signal G_(k) two horizontal periods later than theselection signal G_(k−2) in the second previous row.

Therefore, the odd driver 30 a sequentially activates the selectionsignals G₁, G₃, G₅ . . . in the odd rows every two horizontal periods inthe wake of the activation of the first start pulse SP1 inputted to thefirst-stage unit shift register SR₁ of the odd driver 30 a. On the otherhand, the even driver 30 b sequentially activates the selection signalsG₂, G₄, G₆ . . . in the even rows every two horizontal periods in thewake of the activation of the second start pulse SP2 inputted to thefirst-stage unit shift register SR₂ of the even driver 30 b.

Since the second start pulse SP2 is delayed in phase by one horizontalperiod with respect to the first start pulse SP1, the even driver 30 bstarts its operation one horizontal period later than the odd driver 30a. Therefore, in the gate line drive circuit 30 composed of the odddriver 30 a and the even driver 30 b as a whole, after the activation ofthe first and second start pulses SP1 and SP2, the selection signals G₁,G₂, G₃, G₄ . . . are activated in this order every one horizontalperiod, and the gate lines GL₁, GL₂, GL₃, GL₄ . . . are sequentiallyselected.

As described above, although the selection signal G_(k+1) in the nextrow may be inputted to the reset terminal RST of the unit shift registerSR_(k) according to this embodiment, it is preferable that the selectionsignal G_(k+2) in the second next row is inputted. Hereinafter, thereason will be described.

In order to enhance the resolution of the display panel in the displaydevice having the gate line drive circuit using the shift register, itis necessary to increase the frequency of the clock signal to drive theshift register and speed up the operation of the shift register.However, as the frequency of the clock signal increases, its pulse widthnarrows and an operation margin of the shift register decreases.Consequently, the pulse width of the clock signal is widely set as muchas possible to prevent the margin from decreasing. That is, the intervalbetween the active periods of the clock signals (the interval betweenthe times t₂ and t₃ in FIG. 4, for example) is set to be considerablyshort.

When the interval between the active periods of the clock signals isconsiderably short, the level of the selection signal G_(k+1) in thenext row sometimes starts rising before the level of the selectionsignal G_(k) of the unit shift register SR_(k) becomes sufficiently lowbecause it takes a certain time to discharge the output terminal OUT. Inthis case, when the selection signal G_(k+1) in the next row has beeninputted to the reset terminal RST, the transistor Q5 is turned onbefore the output terminal OUT is sufficiently discharged, so that thelevel of the node N3 falls. In this case, the resistance value of thetransistor Q7 becomes high, and the level of the node N2 rises, so thatthe transistor Q4 is turned on, the level of the node N1 falls, and theresistance value of the transistor Q1 rises. As a result, the fallingspeed of the selection signal G_(k) (discharge speed of the outputterminal OUT) problematically decreases (this problem arisesconspicuously in the following fourth variation of the firstembodiment).

In order to solve the problem, it is considered that the on-resistancevalue of the transistor Q2 is set low, and the output terminal OUT isimmediately discharged in response to the rise of the selection signalG_(k+1) in the next row. However, in order to lower the on-resistance ofthe transistor Q2, it is necessary to increase its gate width, which isnot preferable because a circuit area increases.

In this respect, when the selection signal G_(k+2) in the second nextrow is inputted to the reset terminal RST, a margin about one horizontalperiod can be secured from the start of the discharge of the outputterminal OUT until the start of the discharge of the node N3, so thatthe above problem can be avoided.

Next, a description will be made of the first dummy stage SRD1 providedin the odd driver 30 a and the second dummy stage SRD2 provided in theeven driver 30 b.

FIG. 5 is a circuit diagram of the first dummy stage SRD1. A basiccircuit configuration is about the same as that of the unit shiftregister SR_(k) shown in FIG. 3 except that the selection signal G_(n)of the gate line GL_(n) positioned in the previous row is supplied tothe source of the transistor Q5 to discharge the node N3. In addition,as described above, in the first dummy stage SRD1, the first inputterminal IN1 is supplied with the selection signal G_(n-1), the secondinput terminal IN2 is supplied with the clock signal CLK2, the clockterminal CK is supplied with the clock signal CLK3, and the resetterminal RST is supplied with the clock signal CLK2.

The operation of the first dummy stage SRD1 shown in FIG. 5 is the sameas that of the unit shift register SR_(k) shown in FIG. 3 and describedin the above. However, it is to be noted that since the clock signalCLK2 is supplied to the reset terminal RST (the gate of the transistorQ5), the transistor Q5 is turned on every time the clock signal CLK2 isactivated. Therefore, when the source of the transistor Q5 of the firstdummy stage SRD1 is fixed to the potential VSS like the circuit in FIG.3, the node N3 cannot be boosted with the clock signal CLK2 of thesecond input terminal IN2. Thus, in the first dummy stage SRD1, thesource of the transistor Q5 is put to the H level by the selectionsignal G_(n) only when the node N3 is boosted to prevent the transistorQ5 from being turned on.

FIG. 6 is a circuit diagram of the second dummy stage SRD2. A basiccircuit configuration of the second dummy stage SRD2 is also about thesame as that of the unit shift register SR_(k) shown in FIG. 3 exceptthat the output signal D1 of the first dummy stage SRD1 positioned inthe previous row is supplied to the source of the transistor Q5. Inaddition, as described above, in the second dummy stage SRD2, the firstinput terminal IN1 is supplied with the selection signal G_(n), thesecond input terminal IN2 is supplied with the clock signal CLK3, theclock terminal CK is supplied with the clock signal CLK1, and the resetterminal RST is supplied with the clock signal CLK3.

The operation of the first dummy stage SRD1 shown in FIG. 6 is the sameas that of the unit shift register SR_(k) shown in FIG. 3, and thereason why the output signal D1 of the first dummy stage SRD1 issupplied to the source of the transistor Q5 is the same as the reason ofthe transistor Q5 of the second dummy stage SRD2. That is, in the seconddummy stage SRD2, since the transistor Q5 is turned on every time theclock signal CLK3 is activated, the source of the transistor Q5 is putto the H level by the output signal D1 of the first dummy stage SRD1only when the node N3 is boosted to prevent the transistor Q5 from beingturned on.

As described above, in the gate line drive circuit 30 according to thisembodiment, the plurality of shift registers SR belonging to the odddriver 30 a are cascaded and the plurality of shift registers SRbelonging to the even driver 30 b are cascaded, separately. Even whenthe odd driver 30 a and the even driver 30 b are arranged to sandwichthe liquid crystal array section 10 between them, it is not necessary tosupply the selection signal G to the next-stage unit shift register SRthrough the gate line GL. As a result, the rising speed of the selectionsignal G is prevented from decreasing due to a resistance component anda capacitance component of the gate line GL.

While the first and second dummy stages SRD1 and SRD2 are used as meansfor putting the last stages (unit shift register SR_(n-1) and SR_(n)) ofthe odd driver 30 a and the even driver 30 b to the reset state,respectively in the above description, first and second end pulses EP1and EP2 corresponding to the output signals D1 and D2 may be externallysupplied to the reset terminals RST of the unit shift register SR_(n-1)and SR_(n), respectively.

FIG. 7 is a timing chart showing an operation when the gate line drivecircuit 30 according to this embodiment is controlled with the endpulses. The first end pulse EP1 is a pulse signal activated onehorizontal period later than the selection signal G_(n) in the last row,and the second end pulse EP2 is a pulse signal activated further onehorizontal period later. When the first end pulse EP1 is supplied to thereset terminal RST of the unit shift register SR_(n-1), and the secondend pulse EP2 is supplied to the reset terminal of the unit shiftregister SR_(n), the same operation as that of the gate line drivecircuit 30 in FIG. 2 can be implemented without providing the first andsecond dummy stages SRD1 and SRD2.

In this case, although it is not necessary to provide the first andsecond dummy stages SRD1 and SRD2 in the gate line drive circuit 30, itis necessary to separately provide a generation circuit of the first andsecond end pulses EP1 and EP2 instead.

[First Variation]

The transistor Q6 of the pull-down drive circuit 22 serves as the loadelement of the inverter in the unit shift register SR_(k) in FIG. 3. Theload element of the inverter may be any kind as long as it can keep thenode N2 at the H level while the gate line GL_(k) is not selected.Therefore, any kind of current drive element such as a constant currentelement and a resistor element may be used instead of the transistor Q6.

While the constant high side power supply potential VDD2 is supplied tothe gate of the transistor Q6 in FIG. 3, the clock signal CLK3 (or CLK2)having the same phase as that of the selection signal G_(k+2) in thesecond next row (or selection signal G_(k+1) in the next row) suppliedto the reset terminal RST may be supplied instead. When the unit shiftregister SR_(k) activates the selection signal G_(k), the transistor Q7is in the on state during four horizontal periods (time t₁ to time t₉ inFIG. 4) (or three horizontal periods). While a through current flowsthrough the transistors Q6 and Q7 for those periods in the circuit inFIG. 3, the transistor Q6 is in the off state for three-fourths(two-thirds) periods thereof when the clock signal CLK3 (or CLK2) issupplied to the gate of the transistor Q7, so that the through currentcan be reduced to one-fourth (one-third). Alternatively, both of thegate and drain of the transistor Q6 may be supplied with the clocksignal CLK3 (or CLK2) having the same phase as that of the selectionsignal G_(k+2) in the second next row (or the selection signal G_(k+1)in the next row) supplied to the reset terminal RST.

This variation can be applied to the following all embodiments and theirvariations.

[Second Variation]

FIG. 8 is a circuit diagram of a unit shift register SR_(k) according toa second variation of the first embodiment. The unit shift registerSR_(k) has a transistor Q11 in the pull-down drive circuit 22 withrespect to the circuit in FIG. 3. The transistor Q11 has a gateconnected to the first input terminal IN1, and is connected between thenode N2 and the first power supply terminal S1. In addition, thetransistor Q11 is set such that its on-resistance is sufficiently lowerthan that of the transistor Q6.

In the circuit in FIG. 8, at the point when the selection signal G_(k−2)in the second previous row is activated and the transistor Q8 startscharging the node N3, the transistor Q9 is in the on state. Thetransistor Q9 is turned off when the node N3 is further charged and thetransistor Q7 is turned on and then the node N2 becomes the L level.Therefore, the on-resistance of the transistor Q8 has to be sufficientlylower than that of the transistor Q9.

Meanwhile, in the unit shift register SR_(k) in FIG. 8, the transistorQ11 is turned on at the point when the selection signal G_(k−2) in thesecond previous row is activated, and puts the node N2 to the L level.Therefore, the transistor Q9 is turned off almost as soon as thetransistor Q8 is turned on, and charges the node N3 in that state.Therefore, the node N3 can be precharged regardless of the on-resistancevalues of the transistors Q8 and Q9. That is, since the output of thepull-down drive circuit 22 can be inverted without increasing thedriving capability of the transistor Q8, the parasitic capacitancecaused by the source of the transistor Q8 connected to the node N3 canbe small. Therefore, the boost width of the node N3 by the transistorQ10 (MOS capacitor element) can be larger.

[Third Variation]

FIG. 9 is a circuit diagram of a unit shift register SR_(k) according toa third variation of the first embodiment. In this unit shift registerSR_(k), the source of the transistor Q9 in the pull-up drive circuit 21is connected to the first input terminal IN1 with respect to the circuitin FIG. 3. That is, the source of the transistor Q9 is supplied with theselection signal G_(k−2) in the second previous row.

In the unit shift register SR_(k) in FIG. 9, when the selection signalG_(k−2) in the second previous row is activated, the transistor Q9 isturned off because its source potential increases. That is, thetransistor Q8 is turned off almost as soon as the transistor Q8 isturned on, and charges the node N3 in that state. Therefore, the node N3can be precharged regardless of the on-resistance values of thetransistors Q8 and Q9. This makes circuit designing easy. In addition,there is no through current flowing from the second power supplyterminal S2 to the first power supply terminal S1 through thetransistors Q8 and Q9, so that power consumption can be reduced.

[Fourth Variation]

FIG. 10 is a circuit diagram of a unit shift register SR_(k) accordingto a fourth variation of the first embodiment. In this unit shiftregister SR_(k), the input end of the inverter in the pull-down drivecircuit 22 is connected to the node N1, and a transistor Q12 which isconnected between the node N1 and the first power supply terminal S1 isprovided, with respect to the circuit in FIG. 3. The gate of thetransistor Q12 is connected to the reset terminal RST.

An operation of the unit shift register SR_(k) in FIG. 10 is almost thesame as the operation (FIG. 4) of the circuit in FIG. 3. However, it isto be noted that since the transistor Q5 cannot discharge the input endof the pull-down drive circuit 22 (inverter), the transistor Q12 isprovided instead to discharge it and to put the unit shift registerSR_(k) into the reset state.

According to this variation, since the input end of the pull-down drivecircuit 22 (inverter) is separated from the node N3, the parasiticcapacitance of the node N3 decreases as compared with the circuit inFIG. 3. Therefore, the boost width of the node N3 by the transistor Q10(MOS capacitor element) can be larger.

However, it is to be noted that since the parasitic capacitance of thenode N1 increases, the boost width of the node N1 by the capacitorelement C1 is smaller than that of the circuit in FIG. 3. In whichconfiguration in FIG. 3 or FIG. 10 the rising speed of the selectionsignal G_(k) is higher depends on the conditions such as the transistordimension and circuit layout. The one which is higher in speed of theselection signal G_(k) is to be selected based on those conditions.Furthermore, the above first to third variations may be applied to thisvariation.

[Fifth Variation]

FIG. 11 is a circuit diagram showing a unit shift register according tothe first embodiment. According to this variation, an output terminal VTof a voltage generation circuit 33 to generate a predetermined potentialVDD4 is connected to the drain of the transistor Q3, with respect to theunit shift register SR_(k) according to the first embodiment. While thisvariation may be applied to any unit shift register SR_(k) in the abovefirst to fourth variations, FIG. 11 shows a case where this variation isapplied to the unit shift register SR_(k) in FIG. 3.

The voltage generation circuit 33 generates the potential VDD4 higherthan the H level (VDD) of the clock signals CLK1 to CLK3. The voltagegeneration circuit 33 includes a charge pump circuit driven by the givenclock signal to generate the output potential VDD4 higher than the powersupply potential VDD3 by boosting the predetermined power supplypotential VDD3 (FIG. 11 shows a fourth power supply terminal S4 suppliedwith the power supply potential VDD3, and a clock input terminal CKTsupplied with the clock signal CLK1 representatively). In addition, aconfiguration example of the charge pump circuit is specificallydisclosed in FIGS. 17 to 20 in Japanese Patent Application Laid-Open No.2007-257813, for example.

It is assumed that the output potential VDD4 of the voltage generationcircuit 33 is higher than the potential V3 b (formula (2)) of the nodeN3 after boosted by the transistor Q10 (MOS capacitor element) by thethreshold voltage Vth of the transistor Q3 or more. In this case, sincethe transistor Q3 also operates in the saturated region to charge thenode N1 when the node N3 is boosted, a potential V1 a of the node N1after boosted is as follows.V1a=V3b−Vth  (3)In general, since the potential VDD is set to be sufficiently higherthan the threshold voltage Vth of the transistor, the potential V3 b ishigher than VDD by Vth or more. That is, the following relationship isestablished.V3b=2×VDD−Vth>VDD+Vth  (4)Based on the formulas (3) and (4), the following relationship isprovided.V1a>VDD  (5)That is, the configuration in FIG. 11 increases the gate voltage of thetransistor Q1 and enhances the driving capability of the unit shiftregister SR_(k) as compared with the circuit in FIG. 3.

Second Embodiment

According to a second embodiment, a shift register capable of changing ashift direction of a signal is applied to the present invention. A gateline drive circuit 30 using such shift register is capable ofbidirectional scanning. A direction from head to bottom (order of gatelines GL₁, GL₂, GL₃ . . . ) is defined as a “forward direction”, and adirection from bottom to head (order of gate lines GL_(n), GL_(n-1),GL_(n-2) . . . ) is defined as a “backward direction”.

FIG. 12 is a drawing showing a configuration of the gate line drivecircuit 30 according to the second embodiment. This gate line drivecircuit 30 is also composed of an odd driver 30 a having cascaded unitshift registers SR₁, SR₃, SR₅ . . . to drive the gate lines GL₁, GL₃,GL₅ . . . in the odd rows, and an even driver 30 b having cascaded unitshift registers SR₂, SR₄, SR₆ . . . to drive the gate lines GL₂, GL₄,GL₆ . . . in the even rows.

In addition, control signals (first and second voltage signals Vn and Vras will be described below) to determine the shift direction of thesignal (scanning direction) are supplied to each unit shift register SRaccording to this embodiment, and each unit shift register SR hasterminals (first and second voltage signals Vn and Vr) to receive them,but they are omitted in FIG. 12.

As shown in FIG. 12, the unit shift register SR in this embodiment hasfirst to fourth input terminals IN1 to IN4, an output terminal OUT, aclock terminal CK, a first voltage signal terminal T1, and a secondvoltage signal terminal T2. The clock terminal CK of the unit shiftregister SR is supplied with one of the clock signals CLK1 to CLK3 bythe same rule as in FIG. 2. That is, the clock signal CLK1 is suppliedto the unit shift registers SR₁, SR₄, SR₇ . . . in [3m−2]th rows, theclock signal CLK2 is supplied to the unit shift registers SR₂, SR₅, SR₈. . . in [3m−1]th rows, and the clock signal CLK3 is supplied to theunit shift registers SR₃, SR₆, SR₉ . . . in [3m]th rows.

While the clock signal supplied to the clock terminal CK of the unitshift register SR_(n) (the last stage of the even driver 30 b) in thelast nth row varies depending on the number of scanning lines of thedisplay device, the clock signal CLK2 is supplied to the clock terminalCK of the unit shift register SR_(n) in the example shown in FIG. 12.

A clock signal generator 31 in this embodiment can change the order(phase relationship) of activation of the clock signals CLK1, CLK2, andCLK3 to be outputted, based on the shift direction of the signal, by aswitch, a program, or changing wiring connection. More specifically, theclock signals CLK1 to CLK3 are activated in the order of CLK1, CLK2,CLK3, CLK1 . . . in the case of the forward shift and activated in theorder of CLK3, CLK2, CLK1, CLK3 . . . in the case of the backward shift.Therefore, the clock terminals CK of the shift registers SR₁, SR₂, SR₃ .. . are activated in this order at the time of forward shift, andactivated in reverse order at the time of backward shift.

The means for changing the activation order of the clock signals CLK1 toCLK3 implemented by changing the wiring connection is effective when theshift direction is fixed to one direction before the production of theelectro-optical device. In addition, the means using the switch or theprogram is effective when the shift direction is fixed to one directionafter the production of the electro-optical device or when the shiftdirection can be changed while the electro-optical device is used.

According to this embodiment, an operation of the gate line drivecircuit 30 is controlled with first to fourth control pulses STn1, STn2,STr1, and STr2 functioning as start pulses and end pulses withoutproviding a dummy stage unit shift register. The first to fourth controlpulses STn1, STn2, STr1, and STr2 are generated in a start/end pulsegenerator 34. As a matter of course, a dummy stage to output a signalcorresponding to the end pulse may be provided and in this case, thefirst to fourth control pulses STn1, STn2, STr1, and STr2 only have tofunction as the start pulses.

In the case of the forward shift, the first and second control pulsesSTn1 and STn2 function as the start pulses, and the third and fourthcontrol pulses STr1 and STr2 function as the end pulses. When the firstand second control pulses STn1 and STn2 function as the start pulses,the second control pulse STn2 is activated one horizontal period laterthan the first control pulse STn1 (they behave like the first and secondstart pulses SP1 and SP2 respectively in FIG. 7). In addition, when thethird and fourth control pulses STr1 and STr2 function as the endpulses, the third control pulse STr1 is activated one horizontal periodlater than the fourth control pulse STr2 (they behave like the first andsecond end pulses EP1 and EP2 respectively in FIG. 7).

On the other hand, in the case of the backward shift, the third andfourth control pulses STr1 and STr2 function as the start pulses, andthe first and second control pulses STn1 and STn2 function as the endpulses. Here, it is to be noted that their phase relationship isopposite to that of the forward shift. More specifically, when the firstand second control pulses STn1 and STn2 function as the end pulses, thefirst control pulse STn1 is activated one horizontal period later thanthe second control pulse STn2. In addition, when the third and fourthcontrol pulses STr1 and STr2 function as the start pulses, the fourthcontrol pulse STr2 is activated one horizontal period later than thethird control pulse STr1.

When the bidirectional shift register is provided in the gate line drivecircuit 30 of the image display device like in this embodiment, thestart pulse is activated in response to the head of each frame period ofan image signal. The end pulse is activated in response to the end ofeach frame period.

The first and second input terminals IN1 and IN2 of the unit shiftregister SR₁ in the first row (the first stage of the odd driver 30 a)is supplied with the first and second control pulses STn1 and STn2,respectively. In addition, the first input terminals IN1 of the unitshift register SR₂ in the second row (the first stage of the even driver30 b) is supplied with the second control pulse STn2. In addition, itssecond input terminal IN2 is supplied with the clock signal CLK1 whichis activated one horizontal period later than the second control pulseSTn2 at the time of forward shift.

In the unit shift register SR_(k) on and after the third row, the firstinput terminal IN1 is connected to the output terminal OUT of the unitshift register SR_(k−2) positioned in the second previous row thereof(previous stage in the odd driver 30 a or even driver 30 b). Inaddition, the second input terminal IN2 is supplied with the clocksignal delayed in phase by one horizontal period with respect to theselection signal G_(k−2) in the second previous row at the time offorward shift (that is, the clock signal having the same phase as thatof the selection signal G_(k−1) in the previous row (the same as that ofthe clock terminal CK of the unit shift register SR_(k−1) in theprevious row)). The clock signal of the second input terminal IN2 isadvanced in phase by one horizontal period with respect to the one to besupplied to the clock terminal CK at the time of forward shift.

On the other hand, the third and fourth input terminals IN3 and IN4 ofthe unit shift register SR_(n) in the last row (the last stage of theeven driver 30 b) are supplied with the third and fourth control pulsesSTr1 and STr2, respectively. In addition, the third input terminal IN3of the unit shift register SR_(n-1) in the next-to-last row (the laststage of the odd driver 30 a) is supplied with the fourth control pulseSTr2. In addition, its fourth input terminal IN4 is supplied with theclock signal CLK2 which is activated one horizontal period later thanthe fourth control pulse STr2 at the time of backward shift.

In the unit shift register SR_(k) on and before the third row from thelast, the first input terminal IN1 is connected to the output terminalOUT of the unit shift register SR_(k+2) positioned in the second nextrow thereof (the next stage in the odd driver 30 a or even driver 30 b).In addition, the fourth input terminal IN4 is supplied with the clocksignal delayed in phase by one horizontal period with respect to theselection signal G_(k+2) in the second next row at the time of backwardshift (that is, the clock signal having the same phase as that of theselection signal G_(k+1) in the next row (the same as the that of theclock terminal CK of the unit shift register SR_(k−1) in the previousrow)). The clock signal of the fourth input terminal IN4 is advanced inphase by one horizontal period with respect to the one supplied to theclock terminal CK at the time of backward shift.

Thus, the clock terminal CK of the unit shift register SR_(k) issupplied with the clock signal which is delayed in phase by onehorizontal period with respect to the clock signal of the second inputterminal IN2 at the time of forward shift, and with the clock signalwhich is delayed in phase by one horizontal period with respect to theclock signal of the fourth input terminal IN4 at the time of backwardshift. This switching is implemented by a clock signal generator 31 suchthat the order of the activation of the clock signals CLK1 to CLK3 isreversed between the forward shift and the backward shift as describedabove.

FIG. 13 is a circuit diagram of the unit shift register SR_(k) accordingto this embodiment. In this unit shift register SR_(k), switchingcircuits 24 and 25 to switch the shift direction of the signal areprovided, with respect to the circuit in FIG. 3.

The unit shift register SR_(k) according to this embodiment has thefirst and second voltage signals Vn and Vr supplied with the first andsecond voltage signals Vn and Vr serving as control signals to determinethe shift direction (scanning direction) of the signal. When the forwardshift is performed in the unit shift register SR_(k), the first voltagesignal Vn is set to the active level (H level), and the second voltagesignal Vr is set to the inactive level (L level). In addition, when thebackward shift is performed in the unit shift register SR_(k), the firstvoltage signal Vn is set to the L level, and the second voltage signalVr is set to the H level.

In the circuit in FIG. 3, the gate (first input terminal IN1) of thetransistor Q8 is supplied with the selection signal G_(k−2) in thesecond previous row, and the gate (reset terminal RST) of the transistorQ5 is supplied with the selection signal G_(k+2) in the second next row,while in the circuit in FIG. 13, the switching circuit 24 can switch thetwo signals based on the levels of the first and second voltage signalsVn and Vr. That is, the switching circuit 24 connects the gates of thetransistors Q5 and Q8 to the first and third input terminals IN1 andIN3, respectively, and can switch the connection.

Similarly, in the circuit in FIG. 3, the drain/source of the transistorQ10 is fixedly connected to the second input terminal IN2, while in thecircuit in FIG. 13, the switching circuit 25 can switch the connectionof the drain/source of the transistor Q10 between the second inputterminal IN2 and the fourth input terminal IN4 based on the levels ofthe first and second voltage signals Vn and Vr.

As shown in FIG. 13, the switching circuit 24 is composed of transistorsQ15 r, Q15 n, Q16 r, Q16 n, Q17 r, and Q17 n.

In addition, the switching circuit 24 has input ends of the first andthird input terminals IN1 and IN3 receiving the selection signal G_(k−2)in the second previous row and the selection signal G_(k+2) in thesecond next row, respectively, and two output ends. The two output endsof the switching circuit 24 are defined as a “node N6” and a “node N7”,respectively. Here, the gate of the transistor Q8 is connected to thenode N6, and the gate of the transistor Q5 is connected to the node N7.The node N6 corresponds to the first input terminal IN1 of the circuitin FIG. 3, and the node N7 corresponds to the reset terminal RST of thecircuit in FIG. 3. That is, the switching circuit 24 performs the switchoperation such that either one of the selection signal G_(k−2) in thesecond previous rows or selection signal G_(k+2) in the second next rowis supplied to either one of the first input terminal IN1 in FIG. 3(node N6) or the reset terminal RST in FIG. 3 (node N7).

As shown in FIG. 13, the transistor Q15 n is connected between the firstinput terminal IN1 and the node N6, and the transistor Q15 r isconnected between the third input terminal IN3 and the node N6. Whennodes connected to the gates of the transistors Q15 n and Q15 r aredefined as a “node N8” and a “node N9”, respectively, the transistor Q17n is connected between the node N8 and the first voltage signal terminalT1, and the transistor Q17 n is connected between the node N9 and thesecond voltage signal terminal T2. The gates of the transistors Q17 nand Q17 r are both connected to the second power supply terminal S2.

In addition, the transistor Q16 n is connected between the third inputterminal IN3 and the node N7, and its gate is connected to the firstvoltage signal terminal T1. The transistor Q16 r is connected betweenthe third input terminal IN3 and the node N7, and its gate is connectedto the second voltage signal terminal T2.

The switching circuit 25 is composed of transistors Q18 r, Q18 n, Q19 r,and Q19 n. The switching circuit 25 has the second and fourth inputterminals IN2 and IN4 supplied with predetermined clock signals as itsinput ends, and has the gate of the transistor Q10 as its output end.

As shown in FIG. 13, a node connected to the gate of the transistor Q10is defined as a “node N10”, the transistor Q18 n is connected betweenthe node N10 and the second input terminal IN2, and the transistor 18 ris connected between the node N10 and the fourth input terminal IN4. Inaddition, when nodes connected to the gates of the transistors Q18 n andQ18 r are defined a “node N11” and a “node N12”, respectively, thetransistor Q19 n is connected between the node N11 and the first voltagesignal terminal T1, and the transistor Q19 r is connected between thenode N12 and the second voltage signal terminal T2. The gates of thetransistors Q19 r and Q19 n are both connected to the second powersupply terminal S2.

A description will be made of operations of the switching circuits 24and 25 provided in the unit shift register SR in FIG. 13, hereinafter.Here, it is assumed that the potentials of the H levels of the first andsecond voltage signals Vn and Vr, and the selection signal G_(k−2) inthe second previous row and the selection signal G_(k+2) in the secondnext row (that is, the H level of the clock signals CLK1 to CLK3) areall VDD. In addition, high side power supply potentials VDD1 and VDD2are also equal to VDD.

When the first voltage signal Vn is at the H level (VDD) and the secondvoltage signal Vr is at the L level (VSS), the transistor Q16 n is inthe on state and the transistor Q16 r is in the off state in theswitching circuit 24. In addition, since the gate potentials of thetransistors Q17 n and Q17 r are fixed to the H level of VDD, both are inthe on state. When it is assumed that the selection signal G_(k) of theunit shift register SR_(k), the selection signals G_(k−2) in the secondprevious row and the selection signals G_(k+2) in the second next roware all at the L level, the node N8 is at the H level of the potentialVDD−Vth, and the node N9 is at the L level of the potential VSS.Therefore, the transistor Q15 n is in the on state, and the transistorQ15 r is in the off state.

Therefore, the selection signal G_(k−2) inputted from the secondprevious row to the first input terminal IN1 is supplied to the node N6,and the selection signal G_(k+2) inputted from the second next row tothe second input terminal IN2 is supplied to the node N7.

In the switching circuit 25, since the gate potentials of thetransistors Q19 n and Q19 r are fixed to the H level of VDD, they areboth in the on state. Therefore, when the first voltage signal Vn is atthe H level (VDD) and the second voltage signal Vr is at the L level(VSS), the node N11 is at the H level of the potential VDD−Vth, and thenode N12 is at the L level of the potential VSS. Thus, the transistorQ18 n is in the on state and the transistor Q18 r is in the off state,and the node N10 is supplied with the clock signal of the second inputterminal IN2 (the clock signal CLK3 in FIG. 13).

In this case, the unit shift register SR_(k) in FIG. 13 is equivalent tothat in FIG. 3. Therefore, the selection signal G can be shifted fromthe previous row to the subsequent row (forward direction) in the gateline drive circuit 30 composed of the odd driver 30 a and the evendriver 30 b having the plurality of cascaded unit shift registers SR_(k)in FIG. 13.

In addition, in the case of the forward shift, the first and secondcontrol pulses STn1 and STn2 function as the start pulses, and at thistime, the second control pulse STn2 is activated one horizontal periodlater than the first control pulse STn1. Therefore, the even driver 30 bto which the first row (gate line GL₁) does not belong starts theoperation later than the even driver 30 b to which the first row belongsby one line scanning period. Thus, the selection signals G₁, G₂, G₃ . .. are activated in this order in the gate line drive circuit 30.

In the switching circuit 24 in FIG. 13, when the selection signalG_(k−2) in the second previous row rises and the potential of the nodeN6 rises, the node N8 is boosted due to coupling through the capacitancebetween the gate and the channel of the transistor Q15 n. At this time,since the transistor Q17 is turned off, the node N8 rises to the levelhigh enough to operate the transistor Q25 n in the unsaturated region.Therefore, the potential of the node N6 becomes VDD which is the same asthe H level of the selection signal G_(k−2) in the second previous row.That is, the selection signal G_(k−2) in the second previous row istransmitted to the node N6 without losing the threshold voltage of thetransistor Q15 n.

Similarly, in the switching circuit 25, when the clock signal (clocksignal CLK3) of the second input terminal IN2 rises and the potential ofthe node N10 rises, the node N11 is boosted due to coupling through thecapacitance between the gate and the channel of the transistor Q18 n. Atthis time, since the transistor Q19 n is turned off, the node N11 risesto the level high enough to operate the transistor Q18 n in theunsaturated region. Therefore, the potential of the node N10 becomes VDDwhich is the same as the H level of the clock signal CLK3. That is, theclock signal of the second input terminal IN2 is transmitted to the nodeN10 without losing the threshold voltage of the transistor Q18 n.

On the other hand, when the first voltage signal Vn is at the L level(VSS), and the second voltage signal Vr is at the H level (VDD), thetransistor Q16 n is in the off state and the transistor Q16 r is in theon state in the switching circuit 24. In addition, the transistors Q17 nand Q17 r are both in the on state, and when it is assumed that theselection signal G_(k) of the unit shift register SR_(k), the selectionsignal G_(k−2) in the second previous row and the selection signalG_(k+2) in the second next row are all at the L level, the node N8 is atthe L level of the potential VSS, and the node N9 is at the H level ofthe potential VDD−Vth. Therefore, the transistor Q15 n is in the offstate, and the transistor Q15 r is in the on state.

Therefore, the selection signal G_(k−2) inputted from the secondprevious row to the first input terminal IN1 is supplied to the node N7,and the selection signal G_(k+2) from the second next row to the secondinput terminal IN2 is supplied to the node N6. In this case, the unitshift register SR_(k) in FIG. 13 becomes the set state in response tothe activation of the selection signal G_(k+2) in the second next rowand becomes the reset state in response to the activation of theselection signal G_(k−2) in the second previous row.

In addition, in the switching circuit 25, when the first voltage signalVn is at the L level (VSS) and the second voltage signal Vr is at the Hlevel (VDD), the node N11 is at the L level of the potential VSS, andthe node N12 is at the H level of the potential VDD−Vth. Thus, thetransistor Q18 n is in the off state and the transistor Q18 r is in theon state, and the node N10 is supplied with the clock signal of thefourth input terminal IN4 (the clock signal CLK2 in FIG. 13).

As a result, the selection signals are activated in a direction from thesubsequent row to the previous row (backward direction), that is, theselection signals G_(n), G_(n-1), G_(n-2) . . . are activated in thisorder in the gate line drive circuit 30 composed of the odd driver 30 aand the even driver 30 b having the plurality of cascaded unit shiftregisters SR_(k) in FIG. 13.

As a result, the selection signal G can be shifted in the direction fromthe subsequent row to the previous row (backward direction) in the gateline drive circuit 30 composed of the odd driver 30 a and the evendriver 30 b having the plurality of cascaded unit shift registers SR_(k)in FIG. 13.

In the case of the backward shift, the third and fourth control pulsesSTr1 and STr2 function as the start pulses, and at this time, the fourthcontrol pulse STr2 is activated one horizontal period later than thethird control pulse STr1. Therefore, the odd driver 30 a to which thelast row (gate line GL_(n)) does not belong starts the operation laterthan the even driver 30 b to which the last row belongs by one linescanning period. Thus, the selection signals G_(n), G_(n-1), G_(n-2) . .. are activated in this order in the gate line drive circuit 30.

Since the node N9 is boosted by the coupling through the capacitancebetween the gate and the channel of the transistor Q15 r when theselection signal G_(k+2) in the second next row rises in the unit shiftregister SR_(k), the transistor Q15 r operates in the unsaturatedregion. Thus, the selection signal G_(k+2) in the second next row istransmitted to the node N7 without losing the threshold voltage of thetransistor Q15 r.

Similarly, since the node N12 is boosted by the coupling through thecapacitance between the gate and the channel of the transistor Q18 rwhen the clock signal of the fourth input terminal IN4 (clock signalCLK2) rises, the transistor Q18 r operates in the unsaturated region.Thus, the clock signal of the fourth input terminal IN4 is transmittedto the node N10 without losing the threshold voltage of the transistorQ18 r.

In addition, since operations of an output circuit 20, a pull-up drivecircuit 21, and a pull-down drive circuit 22 in FIG. 13 are same asthose in FIG. 3, the same effect as that in the first embodiment can beprovided in the unit shift register SR_(k) according to this embodiment.Here, it is to be noted that since the signal is supplied to the pull-updrive circuit 21 of the unit shift register SR_(k) through the switchingcircuits 24 and 25, the response to the input signal is a little delayedas compared with the circuit in FIG. 3.

In addition, this embodiment can be applied to any unit shift registerSR_(k) in the first embodiment and its variations.

Third Embodiment

According to a third embodiment, the present invention is applied to agate line drive circuit composed of unit shift registers SR capable ofbidirectional scanning without using a switching circuit. Since a signaldelay due to the switching circuit is not generated in each unit shiftregister, the same response speed as that of a usual unit shift register(performing only one directional shift) can be obtained. Such unit shiftregister is disclosed in Japanese Patent Application Laid-Open No.2008-287753 according to the idea of the inventor of the presentinvention.

FIG. 14 is a circuit diagram of a unit shift register SR_(k) accordingto the third embodiment, in which the present invention is applied tothe circuit in FIG. 3 of Japanese Patent Application Laid-Open No.2008-287753. The unit shift register SR_(k) is also composed of anoutput circuit 20, a pull-up drive circuit 21, and a pull-down drivecircuit 22, and the pull-up drive circuit 21 includes a forward pull-updrive circuit 21 n to drive a transistor Q1 at the time of forward shiftand a backward pull-up drive circuit 21 r to drive the transistor Q1 atthe time of backward shift.

The output circuit 20 has the same configuration as that in FIG. 3, andit is composed of the transistor Q1 to put a selection signal G_(k) intothe active state (H level) while a gate line GL_(k) is selected, and atransistor Q2 to keep the selection signal G_(k) at the inactive state(L level) while the gate line GL_(k) is not selected, and a capacitorelement C1 is provided between the gate and the source of the transistorQ1. Here also, a node connected to the gate (control electrode) of thetransistor Q1 is defined as a “node N1”, and a node connected to thegate of the transistor Q2 is defined as a “node N2”.

The pull-up drive circuit 21 is composed of a transistor Q4, the forwardpull-up drive circuit 21 n, and the backward pull-up drive circuit 21 r.The transistor Q4 has a gate connected to the node N2, and is connectedbetween the node N1 and a first power supply terminal S1 similar to FIG.3.

The forward pull-up drive circuit 21 n is composed of followingtransistors Q3 n, Q5 n, and Q8 n to Q10 n. The transistor Q3 n isconnected between the node N1 and a first voltage signal terminal T1 tosupply a first voltage signal Vn to the node N1. Here, a node connectedto the gate of the transistor Q3 n is defined as a “node N3 n”. Thetransistor Q8 n is connected between the node N3 n and the first voltagesignal terminal T1, and its gate is connected to a first input terminalIN1. The transistor Q10 n has a gate connected to the node N3 n and twocurrent electrodes (source and drain) both connected to a second inputterminal IN2. That is, the transistor Q10 n selectively functions as acapacitor element based on the voltage between the node N3 n and thesecond input terminal IN2.

In addition, the transistor Q5 n is connected between the node N3 n andthe first power supply terminal S1, and its gate is connected to a thirdinput terminal IN3. The transistor Q9 n is connected between the node N3n and the first power supply terminal S1 and its gate is connected tothe node N2.

The backward pull-up drive circuit 21 r is composed of followingtransistors Q3 r, Q5 r, and Q8 r to Q10 r. The transistor Q3 r isconnected between the node N1 and a second voltage signal terminal T2 tosupply a second voltage signal Vr to the node N1. Here, a node connectedto the gate of the transistor Q3 r is defined as a “node N3 r”. Thetransistor Q8 r is connected between the node N3 r and the secondvoltage signal terminal T2, and its gate is connected to the third inputterminal IN3. The transistor Q10 r has a gate connected to the node N3 rand two current electrodes (source and drain) both connected to a fourthinput terminal IN4. That is, the transistor Q10 r selectively functionsas a capacitor element based on the voltage between the node N3 r andthe fourth input terminal IN4.

In addition, the transistor Q5 r is connected between the node N3 r andthe first power supply terminal S1, and its gate is connected to thethird input terminal IN3. The transistor Q9 r is connected between thenode N3 r and the first power supply terminal S1, and its gate isconnected to the node N2.

The pull-down drive circuit 22 is composed of following transistors Q6,Q7 n, and Q7 r. The transistor Q6 is connected between the node N2 and athird power supply terminal S3, and its gate is connected to the thirdpower supply terminal S3 (that is, the transistor Q6 isdiode-connected). The transistor Q7 n has a gate connected to the nodeN3 n, and is connected between the node N2 and the first power supplyterminal S1. The transistor Q7 r has a gate connected to the node N3 r,and is connected between the node N2 and the first power supply terminalS1.

The transistors Q7 n and Q7 r are set such that their on-resistances aresufficiently lower than that of the transistor Q6. Therefore, while thenode N2 is at the H level while both of the nodes N3 and N3 r are at thelow level, it becomes the L level when at least one of the nodes N3 nand N3 r becomes the H level. That is, the pull-down drive circuit 22according to this embodiment is a NOR circuit in which the nodes N3 nand N3 r are input ends and the node N2 is an output end.

A description will be made of an operation of the unit shift registerSR_(k) in FIG. 14, hereinafter. At the time of forward shift, the firstvoltage signal Vn is set at the H level (VDD), and the second voltagesignal Vr is set at the L level (VSS). In this case, the first voltagesignal Vn serves as a power supply of the active level, and the forwardpull-up drive circuit 21 n becomes the active state. Since the drains(first voltage signal terminal T1) of the transistors Q3 n and Q8 n arefixed to the H level (VDD), the forward pull-up drive circuit 21 n andthe transistor Q4 configure a circuit equivalent to the pull-up drivecircuit 21 in FIG. 3.

On the other hand, the backward pull-up drive circuit 21 r is notsupplied with the power supply of the active level and it is in theresting state. In this case, an electric charge is not supplied to thenode N1 through the transistor Q3 r. The transistor Q8 r cannot chargethe node N3 r, and a channel is not formed in the transistor Q10 r (MOScapacitor element) and the node N3 r cannot be boosted. Thus, the nodeN3 r is kept at the L level, and the transistors Q3 r and Q7 r are keptin the off state.

As a result, the pull-up drive circuit 21 and the pull-down drivecircuit 22 in FIG. 14 are equivalent to the pull-up drive circuit 21 andpull-down drive circuit 22 in FIG. 3, respectively. Therefore, the unitshift register SR in FIG. 14 can make the forward shift of the selectionsignal by the same operation as that of the circuit in FIG. 3.

At the time of backward shift, the first voltage signal Vn is set at theL level (VSS) and the second voltage signal Vr is set at the H level(VDD). In this case, the forward pull-up drive circuit 21 n is in theresting state, and the backward pull-up drive circuit 21 r is in theactive state. In addition, the node N3 n is kept at the L level, and thetransistors Q3 n and Q7 n are kept in the off state.

As a result, the pull-up drive circuit 21 and the pull-down drivecircuit 22 in FIG. 14 are equivalent to the pull-up drive circuit 21 andthe pull-down drive circuit 22 in FIG. 3, respectively. Here, it is tobe noted that the backward pull-up drive circuit 21 r operates so as toput the unit shift register SR_(k) into the set state according to theselection signal G_(k+2) from the second next row to the third inputterminal IN3, and put it into the reset state according to the selectionsignal G_(k−2) inputted from the second previous row to the first inputterminal IN1. Therefore, the unit shift register SR in FIG. 14 can makebackward shift of the selection signal.

Thus, since the unit shift register SR_(k) according to this embodimentcan implement the bidirectional scanning without using the switchingcircuit, the same response speed as that of the usual unit shiftregister (performing only the one directional shift) can be obtained.

[Variation]

FIG. 15 is a circuit diagram showing a unit shift register SR_(k)according to a variation of the third embodiment. The unit shiftregister SR_(k) is provided by applying the technique of the fourthvariation of the first embodiment, to the circuit in FIG. 14.

That is, the unit shift register SR_(k) in FIG. 15, in which thepull-down drive circuit 22 is the inverter having the node N1 as itsinput end similar to that in FIG. 10, further includes a transistor Q12n connected between the node N1 and the second voltage signal terminalT2, and a transistor Q12 r connected between the node N1 and the firstvoltage signal terminal T1. The gate of the transistor Q12 r isconnected to the third input terminal IN3, and the gate of thetransistor Q12 n is connected to the first input terminal IN1.

For example, at the time of forward shift (the first voltage signal Vnis at the H level, and the second voltage signal Vr is at the L level),the transistor Q5 n cannot discharge the input end of the pull-downdrive circuit 22 (inverter) in the configuration in FIG. 15. Inaddition, a channel is not formed in the transistor Q10 r (MOS capacitorelement), and the node N3 r cannot be boosted, so that the input end ofthe pull-down drive circuit 22 is not discharged by the transistor Q3 r,either. Therefore, the transistor Q12 n is provided instead to dischargeit and to put the unit shift register SR into the reset state.

Similarly, the transistor Q12 r is to discharge the input end of thepull-down drive circuit 22 (inverter) and put the unit shift registerSR_(k) into the reset state at the time of backward shift.

Thus, the transistors Q12 n and Q12 r in FIG. 15 function similarly tothe transistor Q12 in FIG. 10. Here, it is to be noted that while thetransistor Q12 in FIG. 10 only discharges the node N1, the transistorsQ12 n and Q12 r in FIG. 15 also contribute to charging of the node N1.For example, at the time of forward shift, the transistor Q12 r cancharge the node N1 up to VDD−Vth in response to the activation of theselection signal G_(k−2) in the second previous row, and at the time ofbackward shift, the transistor Q12 n can charge the node N1 up toVDD−Vth in response to the activation of the selection signal G_(k+2) inthe second next row.

According to this variation, since the pull-down drive circuit 22 isseparated from the nodes N3 n and N3 r, parasitic capacitances of thenodes N3 n and N3 r are lower than that of the circuit in FIG. 14. Thus,boost widths of the node N3 n and N3 r by the transistors Q10 n and Q10r (MOS capacitor elements) can be larger.

Here, it is to be noted that since the parasitic capacitance of the nodeN1 increases, the boost width of the node N1 by the capacitor element C1is smaller than that of the circuit in FIG. 14. In which configurationin FIG. 14 or FIG. 15 the rising speed of the selection signal G_(k) ishigher depends on the conditions such as the transistor dimension andcircuit layout. Thus, the one having the high-speed selection signalG_(k) may be employed based on those conditions.

Fourth Embodiment

A fourth embodiment proposes a more desirable structure of the MOScapacitor element (transistors Q10, Q10 n, and Q10 r) in the unit shiftregister SR according to the above first to third embodiments.

In general, the a-Si transistor is high in overlap capacitance between agate and a drain. When the a-Si transistor is used as the unit shiftregister SR, AC power consumption increases due to the overlapcapacitance in the MOS capacitor element because a high frequency clocksignal is supplied to the drain of each MOS capacitor element in theabove unit shift register SR.

The overlap capacitance of the transistor is proportional to its gatewidth. In addition, the gate capacitance value of the transistor as thecapacitance value of the MOS capacitor element is proportional to itsgate area, and the gate area is determined by the product of the gatewidth and a gate length. Therefore, in the MOS capacitor element, inorder to obtain the great capacitance value while suppressing the ACpower consumption caused by the overlap capacitance, the gate width isto be narrowed and the gate length is to be increased in the transistorused as the MOS capacitor element. Therefore, the transistor having astructure in which the gate length Lg is longer than the gate width Wgis preferably used.

Here, it is to be noted that as the gate length increases, theon-resistance (channel resistance) of the transistor increases, so thata time required for a boost operation increases. Thus, it is preferablethat the gate width and the gate length are set to appropriate values inview of the boost speed and the power consumption.

When the MOS capacitor element is composed of the transistor, theon-resistance can be lowered and the time required for the boostoperation can be shortened in the transistor by connecting the twocurrent electrodes (drain and source) of the transistor and using themas one terminal of the MOS capacitor element like the transistor Q10shown in FIG. 3, for example. However, since both of the two currentelectrodes contribute to the overlap capacitance, the overlapcapacitance increases and the power consumption increases.

In addition, even when only one of the two current electrodes of thetransistor is used as the terminal of the MOS capacitor element and theother is made to be in the floating state, the transistor can functionas the capacitor element. In this case, since only one current electrodecontributes to the overlap capacitance, the overlap capacitancedecreases while the on-resistance increases.

FIGS. 16A, 16B, and 16C are drawings showing a transistor structure asthe MOS capacitor element according to the fourth embodiment. FIG. 16Ais a top view of the transistor, and FIG. 16B is a cross-sectional viewtaken along a line A-A in FIG. 16A.

As shown in FIGS. 16A and 16B, the transistor is composed of a glasssubstrate 1, a gate electrode 2, a gate insulating film 3, an activelayer (i layer) 4, an ohmic layer (n layer) 6, a source electrode 7, anda drain electrode 8. The gate electrode 2 is formed on the glasssubstrate 1 and the gate insulating film 3 is formed thereon and theohmic layer 6 is formed thereon through the active layer 4. The ohmiclayer 6 is divided into a part of a source region 6 s and a part of adrain region 6 d, and the source electrode 7 and the drain electrode 8are formed on the source region 6 s and the drain region 6 d,respectively.

In the MOS capacitor element according to this embodiment, only thedrain electrode 8 of the two current electrodes is used as a terminal ofthe MOS capacitor element, and the source electrode 7 is made to be inthe floating state. That is, in the MOS capacitor element, the gateelectrode 2 is used as one terminal, and the drain electrode 8 is usedas the other terminal.

Since the source electrode 7 is in the floating state in the MOScapacitor element in FIGS. 16A and 16B, overlap capacitance whichconsumes an AC power exists only at a region overlapped with the gateelectrode 2 and the drain region 6 d (drain electrode 8). Since thedrain region 6 d is formed smaller than the source region 6 s (sourceelectrode 7) in this MOS capacitor element, so that the overlapcapacitance is low. When the dimension of the drain region 6 d is set toa value as minimum as it can be formed in its production process, theoverlap capacitance can be minimum.

In the MOS capacitor element shown in FIGS. 16A and 16B, a partfunctioning as the capacitor element is a region overlapped with theactive layer 4 and the gate electrode 2, that is, a region (channelregion) in which the channel is formed. The area of the channel regionis increased in the MOS capacitor element in FIGS. 16A and 16B when itis roughly a square in shape. As a result, the channel resistancebecomes low, so that the time required for the boost operation isshortened in the MOS capacitor element, and transmission delay is hardlygenerated.

As described above, in the transistor according to this embodiment, thedrain electrode 8 serving as one terminal of the MOS capacitor elementis narrower in width Wd than that Ws of the source electrode 7 in thefloating state. Since the drain electrode 8 is small, the overlapcapacitance is low. In addition, since the source electrode 7 and thegate electrode 2 are large, the channel region can be large, so that thechannel resistance can be small and the great capacitance value can beensured. Therefore, the MOS capacitor element composed of the abovetransistor can be superior in characteristics in view of the boost speedand power consumption.

As the drain electrode 8 can be formed at any position, the drainelectrode 8 may be formed at a corner of the gate electrode 2, forexample. However, the drain electrode 8 is preferably arranged in thecenter of one side of the gate electrode 2 as shown in FIGS. 16A and 16Bso that the channel resistance can be easily lowered. In addition, theplurality of drain electrodes 8 may be provided and each may be arrangedin each center of a pair of opposite sides (two opposed sides) of thegate electrode 2, for example. In addition, each may be arranged in thecenter of each side of the gate electrode 2. Here, it is to be notedthat a gate wiring (a part extending outside from the gate electrode 2)has to be formed based on the position of the drain electrode 8.Although the channel resistance can be further lowered by providing theplurality of drain electrodes 8, it is to be noted that the overlapcapacitance of the gate and the drain increases according to the numberof the drain electrodes 8.

A production method of the transistor in FIGS. 16A and 16B will bedescribed. First, a Cr film is formed to be 0.3 μm in thickness on theglass substrate 1 by sputtering, and it is patterned with a photoresistmask, whereby the gate electrode 2 and the gate wiring are formed. Then,an SiN film is formed to be 0.2 μm in thickness thereon by CVD (ChemicalVapor Deposition), whereby the gate insulating layer 3 is formed.

Then, an i-type amorphous silicon layer serving as the active layer 4 isformed to be 0.1 μm in thickness by CVD, and an n-type amorphous siliconlayer serving as the ohmic layer 6 is formed to be 0.05 μm in thicknessby CVD. Then, those films are patterned, whereby the ohmic layer 6 andthe active layer 4 are formed.

Then, a metal layer such as a Cr layer is formed to be 0.1 μm inthickness by sputtering, and patterned, whereby the source electrode 7and the drain electrode 8 are formed. Then, the ohmic layer 6 betweenthe source electrode 7 and the drain electrode 8 is etched away with thesource electrode 7 and the drain electrode 8 used as a mask. Thus, theohmic layer 6 is divided into the source region 6 s and the drain region6 d.

The source electrode 7 is in the floating state in the transistor as theMOS capacitor element in FIGS. 16A and 16B. When the transistor is usedas the transistor Q10 of the unit shift register SR_(k) in FIG. 3, forexample, the drain electrode 8 is connected to the second input terminalIN2, and the gate electrode 2 is connected to the node N3. In this case,when the clock signal supplied to the second input terminal IN2 becomesthe H level, a positive electric charge is gradually accumulated in thesource electrode 7 due to a leak current from the drain electrode 8.Therefore, when it takes a long time in the operation of the unit shiftregister SR_(k), the potential of the source electrode 7 becomes highand there is a concern that the channel cannot be formed between thegate electrode 2 and the source electrode 7 when the node N3 becomes theH level. Therefore, the source electrode 7 is preferably fixed to thepotential at the L level.

FIG. 17 is an example of a unit shift register SR_(k) using the MOStransistor according to the fourth embodiment. In this unit shiftregister SR_(k), the source (corresponding to the source electrode 7 inFIGS. 16A and 16B) of the transistor Q10 is connected to a first powersupply terminal S1 through a transistor Q20, with respect to the circuitin FIG. 3. The gate of the transistor Q20 is connected to a node N2.Here, a node connected to the source of the transistor Q10 is defined asa “node N13”.

In the configuration in FIG. 17, while the unit shift register SR_(k) isnot selected (period in the reset state), an electric charge enteringthe node N13 due to the leak current of the transistor Q10 is dischargedto the power supply terminal S1 through the transistor Q20, so that thenode N13 is kept at the L level. Therefore, the above problem in thetransistor Q10 can be solved.

In addition, while the unit shift register SR_(k) is selected (period inthe set state), the transistor Q20 is in the off state, so that the nodeN13 is in the floating state. Thus, a through current is prevented frombeing generated in the transistors Q10 and Q20 while the node N3 isboosted.

The MOS capacitor element (transistor Q10) in FIG. 17 can be applied toany unit shift register SR according to the first to third embodimentsand their variations. In addition, while the channel-etch type a-Sitransistor is shown in FIGS. 16A and 16B, this embodiment can be appliedto an etch-stopper type or top-gate type a-Si transistor as a matter ofcourse.

Fifth Embodiment

A fifth embodiment proposes a technique to improve the quality of acaptured image and a displayed image in the electro-optical deviceaccording to the present invention.

For example, through wiring paths of the clock signals CLK1 to CLK3 inthe gate line drive circuit 30 in FIG. 2, the clock signal CLK1outputted from the clock signal generator 31 drives the unit shiftregister SR_(n-1) at first and drives the unit shift register SR_(n-4)(not shown) at last (except for the first and second dummy stages SRD1and SRD2). Similarly, the clock signal CLK2 drives the unit shiftregister SR_(n-3) at first and drives the unit shift register SR_(n) atlast. In addition, the clock signal CLK3 drives the unit shift registerSR_(n-5) (not shown) at first and drives the unit shift registerSR_(n-2) at last. Thus, there is a large difference in wiring length(roughly corresponding to a length of three sides of the display panel)from the clock signal generator 31 between the unit shift registers SRto be driven at first and at last regarding each of the clock signalsCLK1 to CLK3.

In general, the wiring has a parasitic resistance component and aparasitic capacitance component and they increase in proportion to thewiring length. Therefore, as the wiring length of the clock signalincreases, the rising time and the falling time of the clock signalincrease due to the parasitic components of the wiring. This causes anincrease in rising and falling times of the selection signal of the gateline.

This phenomenon also appears in the same clock wiring, and the risingand falling times of the clock signal is delayed with the distance fromthe clock signal generator 31. Therefore, the wiring path in FIG. 2 hasa difference in rising and falling speed of the outputted selectionsignal G between the unit shift registers SR_(n-1), SR_(n-3), andSR_(n-5), and the unit shift registers SR_(n-4), SR_(n-2), and SR_(n).As a result, there is a certain difference in written signal voltagebetween the pixels connected to the gate lines GL_(n-1), GL_(n-3), andGL_(n-5), and pixels connected to the gate lines GL_(n-4), GL_(n-2), andGL_(n).

As can be seen from FIG. 2, the gate lines GL_(n-1), GL_(n-3), andGL_(n-5), and the gate lines GL_(n-4), GL_(n-2), and GL_(n) are allarranged in the vicinity of the last row. In general, when the pluralityof pixels show the same color, it is difficult to recognize thedifference on the display with the naked eye in the case where thepixels are apart from each other even when there is a small differencein signal voltage in the pixels. However, the difference can berecognized when the pixels are close to each other, so that the abovedifference in rising and falling speed of the selection signal G isproblematic.

FIG. 18 is a drawing showing a configuration of a gate line drivecircuit 30 according to the fifth embodiment, in which clock wirings arearranged so that the clock signals CLK1 to CLK3 from the clock signalgenerator 31 can be supplied to the odd driver 30 a and the even driver30 b with almost the same wiring length, with respect to theconfiguration in FIG. 2.

As shown in FIG. 18, each wiring of the clock signals CLK1 to CLK3extending from the clock signal generator 31 is drawn to the upper side(top row) of the liquid crystal array section 10 without being connectedto the unit shift register SR, and branched to the odd driver 30 a sideand the even driver 30 b side. The clock wirings on the odd driver 30 aside are sequentially connected to the unit shift registers SR from thetop (in the order of unit shift registers SR₁, SR₃, and SR₅) at thepositions close to branch points of the clock wirings. Similarly, theclock wirings on the even driver 30 b side are sequentially connected tothe unit shift registers SR from the top (in the order of unit shiftregisters SR₂, SR₄, and SR₆) at the positions close to branch points ofthe clock wirings. As a result, the unit shift registers SR to drive theadjacent rows have almost the same wiring distance from the clock signalgenerator 31.

As a matter of course, although the rising and falling speed of theclock signal decreases according to the wiring distance even in thewiring layout in FIG. 18, the clock signals having almost the samerising and falling speed are supplied to the unit shift registers SRarranged in the adjacent rows. Therefore, there is no difference insignal voltage caused by the difference in rising and falling speed ofthe clock signals between the pixels in the adjacent rows. As a result,the difference on the display is prevented from being recognized withthe naked eye.

In addition, it is assumed that the source driver 40 described in FIG. 1is arranged on the lower side (bottom row side) of the liquid crystalarray section 10 in FIG. 18. Although the wiring layout in FIG. 18 canbe applied to the arrangement in which the source driver 40 is disposedon the upper side (top row side) as shown in FIG. 1, this is notpreferable because the wirings of the clock signals CLK1 to CLK3intersect with the data line DL. When both wirings intersect with eachother, the problem is that as well as causing the load capacitance ofthe data line D1 to increase, both of the wirings are capacitivelycoupled at the intersection point, and the level change of the clocksignals CLK1 to CLK2 having large amplitude is problematicallysuperimposed on the fine signal of the data line DL as a noise.

Sixth Embodiment

In the unit shift register SR shown in the first to fourth embodiments,the transistor Q3 is operated in the unsaturated region by boosting thegate (node N3) of the transistor Q3 (or Q3 n or Q3 r) which charges thegate (node N1) of the transistor Q1, with the transistor Q10 (or Q10 nor Q10 r) serving as the MOS capacitor element. Thus, the node N1 ischarged (precharged) to the potential VDD.

However, since it is necessary to ensure a relatively large gate area inthe MOS capacitor element, there is a problem that its area increases.Thus, according to this embodiment, a unit shift register SR in whichthe gate of the transistor Q3 is boosted when the node N1 is charged isimplemented without using the MOS capacitor element. Thus, the displaydevice can be smaller in area.

FIG. 19 is a circuit diagram showing a configuration of the unit shiftregister SR according to a sixth embodiment. The unit shift registerSR_(k) in the kth stage will be representatively described. As shown inFIG. 19, the unit shift register SR_(k) has also a first input terminalIN1, a second input terminal IN2, an output terminal OUT, a clockterminal CK, and a reset terminal RST similar to the circuit in FIG. 3,and it can be used as the unit shift registers SR₁ to SR_(n) in FIG. 2,for example.

The unit shift register SR_(k) according to this embodiment is alsocomposed of an output circuit 20, a pull-up drive circuit 21 and apull-down drive circuit 22. The output circuit 20 includes a transistorQ1 (output pull-up transistor) to put the selection signal G_(k) to theactive state (H level) while the gate line GL_(k) is selected, andtransistors Q2 and Q25 (output pull-down transistors) to keep theselection signal G_(k) in the inactive state (L level) while the gateline GL_(k) is not selected.

The transistor Q1 is connected between the output terminal OUT and theclock terminal CK, and when the clock signal inputted to the clockterminal CK is supplied to the output terminal OUT, the selection signalG_(k) is activated. In addition, the transistors Q2 and Q25 areconnected between the output terminal OUT and a first power supplyterminal S1 and when the output terminal OUT is discharged to thepotential VSS, the selection signal G_(k) is kept at the inactive level.Here, a node connected to the gate (control electrode) of the transistorQ1 is defined as a “node N1”. The gate of the transistor Q2 is connectedto an output end (defined as a “node N2”) of the pull-down drive circuit22 as will be described below.

A capacitor element C1 is provided between the gate and the source ofthe transistor Q1 (that is, between the output terminal OUT and the nodeN1). The capacitor element C1 is provided to enhance the boost effect ofthe node N1 associated with an increase in level of the output terminalOUT. The capacitor element C1 can be replaced with the transistor Q1when the capacitance between the gate and the channel of the transistorQ1 is sufficiently high, so that it may be omitted in that case.

The pull-up drive circuit 21 drives the transistor Q1 (output pull-uptransistor) and operates so as to put the transistor Q1 to the on statewhile the gate line GL_(k) is selected and to put it to the off statewhile the gate line GL_(k) is not selected. Therefore, the pull-up drivecircuit 21 charges the node N1 (transistor Q1) in response to theactivation of a selection signal G_(k−2) inputted from the secondprevious row to the first input terminal IN1 (or a first or second startpulse SP1 or SP2) and the clock signal inputted to the second inputterminal IN2 (the clock signal CLK3 in the case in FIG. 19), anddischarges the node N1 in response to the activation of a selectionsignal G_(k+2) inputted from the second next row to the reset terminalRST as a reset signal (or output signals D1 or D2 of dummy stages SRD1or SRD2).

In the pull-up drive circuit 21, a transistor Q3 is connected betweenthe node N1 and the second input terminal IN2 to charge the node N1 bysupplying the H level potential VDD of the second input terminal IN2 tothe node N1. In addition, a transistor Q4 is connected between the nodeN1 and the first power supply terminal S1 to discharge the node N1 bysupplying the potential VSS of the first power supply terminal S1 to thenode N1. The gate of the transistor Q4 is connected to the node N2.

A node connected to the gate of the transistor Q3 is defined as a “nodeN3”. The pull-up drive circuit 21 has a transistor Q21 having onecurrent electrode connected to the node N3, and a gate connected to thesecond power supply terminal S2. The other current electrode of thetransistor Q21 is defined as a “node N4”.

A transistor Q8 is connected between a second power supply terminal S2and the node N4, and its gate is connected to the first input terminalIN1. The drain of the transistor Q8 may be connected to the first inputterminal IN1 together with the gate. In addition, transistors Q9 and Q24are connected between the node N4 and the first power supply terminalS1. The gate of the transistor Q9 is connected to the node N2. Inaddition, the gate of the transistor Q24 is defined as a “node N5”.

A transistor Q22 is connected between the node N5 and the second powersupply terminal S2, and its gate is connected to the node N1. Atransistor Q23 is connected between the node N5 and the first powersupply terminal S1, and its gate is connected to the node N2.

The gate of the transistor Q25 of the output circuit 20 described aboveis connected to the node N4 of the pull-up drive circuit 21 as shown inFIG. 19.

The pull-down drive circuit 22 has an inverter composed of transistorsQ6 and Q7 connected between the third power supply terminal S3 and thefirst power supply terminal S1 in series, and the output end of theinverter serves as the output end (node N2) of this pull-down drivecircuit 22. The transistor Q6 is connected between the node N2 and thethird power supply terminal S3, and its gate is connected to the thirdpower supply terminal S3. The transistor Q7 is connected between thenode N2 and the first power supply terminal S1. A node connected to thegate of the transistor Q7 serves as the input end of the inverter andthis is defined as a “node N6”.

The input level of the inverter (level of the node N6) is controlled byan input circuit composed of transistors Q26 and Q27. The transistor Q26is connected between the node N6 and the third power supply terminal S3,and its gate is connected to the first input terminal IN1. Thetransistor Q27 is connected between the node N6 and the first powersupply terminal S1, and its gate is connected to the reset terminal RST.This input circuit puts the node N6 to the H level in response to theactivation of the selection signal G_(k−2) in the second previous row,and puts the node N6 to the L level in response to the activation of theselection signal G_(k+2) in the second next row. Therefore, the outputlevel of the inverter (level of the node N2) becomes the L level inresponse to the activation of the selection signal G_(k−2) in the secondprevious row, and becomes the H level in response to the activation ofthe selection signal G_(k+2) in the second next row.

In addition, the pull-down drive circuit 22 has a transistor Q28 thathas a gate connected to the node N2 and is connected between the node N6and the first power supply terminal S1. The transistor Q28 is turned onwhen the node N2 becomes the H level, and keeps the node N6 at the Llevel of low impedance. The transistor Q28 prevents the node N6 frombecoming a floating state after the input circuit sets the node N6 atthe L level, and functions to prevent an error operation of thepull-down drive circuit 22. In addition, the Q 28 is set such that itson-resistance is sufficiently lower than the transistor Q26.

FIG. 20 is a timing chart to describe an operation of the unit shiftregister SR_(k) according to the sixth embodiment. The operation of theunit shift register SR_(k) will be described with reference to thedrawing hereinafter.

Here, it is assumed that the clock signal CLK1 is inputted to the clockterminal CK of the unit shift register SR_(k). Similar to the firstembodiment, since the second input terminal IN2 is supplied with theclock signal advanced in phase by one horizontal period with respect tothe one supplied to the clock terminal CK, the second input terminal IN2of the unit shift register SR_(k) is supplied with the clock signalCLK3.

As an initial state of the unit shift register SR_(k) (just before atime t₁), it is assumed that the nodes N1, N3, N4, N5, and N6 are set tothe L level (VSS) and the node N2 is set at the H level (VDD−Vth) (resetstate). In this state, since the transistor Q1 is in the off state andthe transistor Q2 is in the on state, the output terminal OUT (selectionsignal G_(k)) is at the L level of low impedance regardless of the levelof the clock signal CLK1.

In addition, just before the time t₁, it is assumed that the clockterminal CK (clock signal CLK1), the first input terminal IN1 (selectionsignal G_(k−2) in the second previous row), the second input terminalIN2 (clock signal CLK3) and the reset terminal RST (selection signalG_(k+2) in the second next row) are all set to the L level in the unitshift register SR_(k). In this case, since the node N2 is at the Hlevel, the node N3 is at the L level (VSS) of low impedance through thetransistors Q9 and Q21 in the on state. In addition, the transistors Q4and Q28 are in the on state, and the nodes N1 and N6 are also at the Llevel (VSS) of low impedance.

From this state, it is assumed that the selection signal G_(k−2) in thesecond previous row (start pulse SP1 in the case of the unit shiftregister SR₁ in the first stage) is activated at the time t₁. Thus, thetransistor Q26 is turned on in the pull-down drive circuit 22. Since theon-resistance value of the transistor Q26 is set sufficiently lower thanthe on-resistance value of the transistor Q28, the level of the node N6increases. Accordingly, the transistor Q7 is turned on and the level ofthe node N2 decreases. Then, the transistor Q28 is turned off and thenode N6 is rapidly charged to the H level (VDD−Vth). The node N2 becomesthe L level potential (roughly VSS) determined by the on-resistanceratio of the transistors Q6 and Q7.

On the other hand, the transistor Q8 is turned on in the pull-up drivecircuit 21. In addition, since the on-resistance value of the transistorQ9 becomes high due to the decrease in level of the node N2, the node N4is charged to the H level (VDD−Vth). The increase in level of the nodeN4 is transmitted to the node N3 through the transistor Q21, and thenode N3 also becomes the H level (VDD−Vth).

When the node N2 becomes the L level, the transistors Q1 and Q2 becomethe off state as described above, but the transistor Q25 of the outputcircuit 20 is turned on because the node N4 becomes the H level, so thatthe output terminal OUT is kept at the L level of low impedance. As willbe described below, since the period during which the transistors Q1 andQ2 are both in the off state is roughly equal to one horizontal period(1H), the transistor Q25 may be omitted when the noise of the gate lineGL_(k) is small in this period.

When the selection signal G_(k−2) in the second previous row becomes theL level at a time t₂, the transistors Q8 and Q26 are turned off, butsince the nodes N3, N4, and N6 are kept in the floating state at the Hlevel, there is no level change in each node of the unit shift registerSR_(k).

When the clock signal CLK3 is activated at a time t₃, the node N1 ischarged through the transistor Q3 in the on state, and the level of thenode N1 increases. At this time, the capacitance is coupled between thesecond input terminal IN2, and the node N1 and the node N3 by theparasitic capacitance of the transistor Q3 (such as the capacitancebetween the gate and the channel, and the overlap capacitance betweenthe gate and the source/drain), so that the level of the node N3increases according to the increase in level of the second inputterminal IN2 and the node N1. When the node N3 increases in level, thetransistor Q21 is turned off (the operation of the transistor Q21 willbe described in detail below), and the node N3 increases in level to behigh enough to operate the transistor Q3 in the unsaturated region.Therefore, the node N1 is charged (precharged) at high speed andimmediately becomes the H level of the potential VDD following the clocksignal CLK3. Accordingly, the transistor Q1 is turned on.

On the other hand, the transistor Q22 is turned on in response to theincrease in level of the node N1. Since the transistor Q23 is in the offstate, the node N5 is charged to the H level. Accordingly, thetransistor Q24 is turned on, and the level of the node N4 decreases.Since the transistor Q21 is turned on in response to the decrease inlevel of the node N4, the node N3 is discharged to the L level almost atthe same time with the node N4. Thus, the transistor Q3 is turned off.In FIG. 19, a time t_(3D) represents the time when the node N3 changesto the L level.

Here, attention is paid to the operation of the transistor Q21 when thenode N1 is precharged. Before the node N1 is precharged, the node N4 isat the H level (VDD−Vth), and the gate voltage of the transistor Q21 isfixed to the VDD (=VDD1), so that the transistor Q21 applies a currentfrom the node N4 to the node N3, and charges the node N3 to the H level(VDD−Vth).

Thus, when the clock signal CLK3 rises and the transistor Q3 startsprecharging the node N1, the node N3 is boosted, so that the node N4side becomes the source of the transistor Q21 based on the potentialrelationship. At this point, since the potential of the node N4 is atVDD−Vth, the voltage between the gate (second power supply terminal S2)and the source (node N4) of the transistor Q21 becomes Vth and thetransistor Q21 becomes a boundary state between the on state and the offstate. Thus, a sub-threshold current is applied to the transistor Q21 ina direction from the node N3 to the node N4. Since the current is verysmall, an electric charge discharged from the node N3 for a short period(from time t₃ to t_(3D)) during which the node N3 is boosted can be assmall as negligible.

Thus, when the node N4 becomes the L level after the node N1 has beenprecharged to the H level (VDD), the transistor Q21 is turned on, andthe current flows from the node N3 to the node N4, and the node N3becomes the L level (VSS). After that, the transistor Q21 is still inthe on state while the node N4 is at the L level, and the node N3 iskept at the L level.

As described above, the transistor Q21 serves as a resistance element totransmit the potential of the node N4 to the node N3 at the stage inwhich the node N4 is at the H level before the precharge of the node N1,and serves as a cut-off element to cut off the transmission between thenode N3 and the node N4 at the stage in which the node N3 has beenboosted at the time of the precharging start of the node N1. Inaddition, at the stage in which the node N1 is further precharged andthe level of the node N4 is decreasing, and at the stage in which thenode N4 is kept at L level after that, the transistor Q21 serves as aresistance element to discharge the electric charge of the node N3 tothe node N4. That is, the transistor Q21 serves as acharging/discharging circuit which charges the node N3 after theselection signal G_(k−2) in the second previous row is activated andbefore the clock signal CLK3 is activated, and then discharges the nodeN3 before the clock signal CLK3 is inactivated.

While the description has been made assuming that the potential (VDD1)supplied to the gate of the transistor Q21 is at the VDD which is thesame potential of the H level of the selection signal G at each stage(the potential of the L level of the clock signals CLK1 to CLK3), thepotential supplied to the gate of the transistor Q21 may be anypotential as long as the node N3 can be charged to the H level when thenode N4 is at the H level, and the current does not flow from the nodeN3 to the node N4 through the transistor Q21 when the node N3 isboosted. Although this potential may be lower then VDD, in this case, itis to be noted that the charging speed of the node N1 by the transistorQ3 decreases because the level of the node N3 after boosted is low.

FIG. 20 is referred to again. While the clock signal CLK3 becomes the Llevel at a time t₄, the node N1 is kept at the H level of the potentialVDD because the transistor Q3 has been already turned off at the timet_(3D).

Thus, when the clock signal CLK1 rises at a time t₅, the level change istransmitted to the output terminal OUT through the transistor Q1 in theon state, and the level of the selection signal G_(k) increases. At thistime, the node N1 is boosted due to the coupling through the capacitorelement C1, and the transistor Q1 is operated in the unsaturated region.Thus, the selection signal G_(k) becomes the H level of the potentialVDD which is the same as the H level of the clock signal CLK1.

When the clock signal CLK1 falls at a time t₆, a current flows from theoutput terminal OUT to the clock terminal CK through the transistor Q1in the on state, and the output OUT is discharged. As a result, theselection signal G_(k) becomes the L level. At this time, the node N1returns to the level (VDD) before boosted by the coupling through thecapacitor element C1.

The clock signal CLK2 is activated at a time t₇, and then the clocksignal CLK2 is inactivated at a time t₈, but there is no level change ineach node of the unit shift register SR_(k) at this time.

When the clock signal CLK3 rises at a time t₉, the selection signalG_(k+2) in the second next row becomes the H level. Thus, the transistorQ27 is turned on and the node N6 becomes the L level in the unit shiftregister SR_(k). Accordingly, the transistor Q7 is turned off and thenode N2 is charged by the transistor Q6 to the H level.

When the node N2 becomes the H level, the transistor Q4 is turned on andthe node N1 is discharged to the L level. Accordingly, the transistor Q1is turned off, but the output terminal OUT is kept at the L level of lowimpedance because the transistor Q2 is turned on. In addition, since thetransistor Q23 is turned on and the transistor Q22 is turned off, thenode N5 is discharged to the L level. Accordingly, the transistor Q24 isturned off, but the transistor Q9 is turned off at this time, so thatthe nodes N4 and N3 are kept at the L level of low impedance. As aresult, the unit shift register SR_(k) returns to the reset state inwhich the transistor Q1 is in the off state and the transistor Q2 is inthe on state.

Then, although the selection signal G_(k+2) in the second next rowreturns to the L level at a time t₁₀, the unit shift register SR_(k) iskept in the reset state until the selection signal G_(k−2) in the secondprevious rows is activated in the next frame period. This is because ahalf latch circuit composed of the transistors Q6, Q7, and Q28 keeps thelevel of the nodes N2 and N6. In addition, since the transistor Q2 is inthe on state during that period, the output terminal OUT is kept at theL level of low impedance.

In the unit shift register SR according to this embodiment, theparasitic capacitance of the transistor Q3 (such as the capacitancebetween the gate and the channel, and the overlap capacitance betweenthe gate and the source/drain) functions as a boosting means forboosting the node N3 when the clock signal CLK3 is activated after theselection signal G_(k−2) in the second previous row has been activated.This boosting means increases the gate potential of the transistor Q3when the node N1 is precharged, and operates the transistor Q3 in theunsaturated region. Thus, it is not necessary to separately provide theMOS capacitor element to boost the node N3. Therefore, the gate linedrive circuit which is smaller in area can be implemented.

Seventh Embodiment

According to a seventh embodiment, the technique according to the sixthembodiment is applied to the shift register capable of changing theshift direction of the signal. FIG. 21 is a circuit diagram showing aconfiguration of a unit shift register SR according to the seventhembodiment. Here also, the unit shift register SR_(k) in the kth stagewill be described representatively. The unit shift register SR_(k) maybe used as the unit shift register SR₁ to the unit shift register SR_(k)in FIG. 12, for example.

Similar to the circuit in FIG. 13, the unit shift register SR_(k) hasfirst to fourth input terminals IN1 to IN4, an output terminal OUT, aclock terminal CK, a first voltage signal terminal T1, and a secondvoltage signal terminal T2. Signals supplied to these terminals are thesame as described in the second embodiment (FIG. 12).

Here, a description will be made assuming that the clock terminal CK issupplied with the clock signal CLK1, the second input terminal IN2 issupplied with the clock signal CLK3, and the fourth input terminal IN4is supplied with the clock signal CLK2 in the unit shift register SR_(k)(corresponding to the unit shift register SR₄ in FIG. 12).

In the unit shift register SR_(k) in FIG. 21, an output circuit 20 hasthe same configuration as that in FIG. 19. That is, the output circuit20 is composed of a transistor Q1 to supply the clock signal CLK1 to theoutput terminal OUT, and transistors Q2 and Q25 to discharge the outputterminal OUT during an unselected period.

Similar to FIG. 19, a pull-down drive circuit 22 is also composed of aninverter composed of transistors Q6 and Q7, an input circuit composed oftransistors Q26 and Q27, and a transistor Q28, and the configuration ofthe input circuit is different from that in FIG. 19. That is, in theunit shift register SR_(k) shown in FIG. 21, the transistor Q26 isconnected between the first voltage signal terminal T1 and a node N6(the gate of the transistor Q7), and its gate is connected to the firstinput terminal IN1. The transistor Q27 is connected between the secondvoltage signal terminal T2 and the node N6, and its gate is connected tothe third input terminal IN3. The transistors Q26 and Q27 are set suchthat their on-resistances are sufficiently lower than that of thetransistor Q28.

Therefore, at the time of forward shift (the first voltage signal Vn isat the H level, and the second voltage signal Vr is at the L level), theinput circuit puts the node N6 to the H level in response to theactivation of the selection signal G_(k−2) in the second previous row,and puts the node N6 to the L level in response to the activation of theselection signal G_(k+2) in the second next row. Meanwhile, at the timeof backward shift (the first voltage signal Vn is at the L level, andthe second voltage signal Vr is at the H level), it puts the node N6 tothe H level in response to the activation of the selection signalG_(k+2) in the second next row, and puts the node N6 to the L level inresponse to the activation of the selection signal G_(k−2) in the secondprevious row.

The gate of the transistor Q2 of the output circuit 20 is connected to anode N2 serving as the output end of the inverter composed of thetransistors Q6 and Q7 (the output end of the pull-down drive circuit 22)similar to FIG. 19.

Meanwhile, in a pull-up drive circuit 21, while a part composed oftransistors Q4, Q9, Q22, Q23, Q24 is the same circuit as in FIG. 19,remaining part is a circuit composed of the following transistors Q3 n,Q3 r, Q21 n, Q21 r, Q29 n, Q29 r, Q8 n, and Q8 r. Similar to the case inFIG. 19, a node connected to drains of the transistors Q24 and Q9 isdefined as a “node N4”. The gate of the transistor Q25 of the outputcircuit 20 is connected to the node N4.

The transistor Q3 n is connected between the second input terminal IN2and the gate (node N1) of the transistor Q1. When a node connected tothe gate of the transistor Q3 n is defined as a “node N3 n”, thetransistor Q21 n is connected between the node N3 n and the node N4, andits gate is connected to the first voltage signal terminal T1. Thetransistor Q29 n is connected between the node N3 n and the firstvoltage signal terminal T1, and its gate is connected to the secondvoltage signal terminal T2. The transistor Q8 n is connected between thefirst voltage signal terminal T1 and the node N4, and its gate isconnected to the first input terminal IN1.

The transistor Q3 r is connected between the fourth input terminal IN4and the node N1. When a node connected to the gate of the transistor Q3r is defined as a “node N3 r”, the transistor Q21 r is connected betweenthe node N3 r and the node N4, and its gate is connected to the secondvoltage signal terminal T2. The transistor Q29 r is connected betweenthe node N3 r and the second voltage signal terminal T2, and its gate isconnected to the first voltage signal terminal T1. The transistor Q8 ris connected between the second voltage signal terminal T2 and the nodeN4, and its gate is connected to the third input terminal IN3.

At the time of forward shift, since the first voltage signal Vn is atthe H level (VDD) and the second voltage signal Vr is at the L level(VSS), in the pull-up drive circuit 21, the transistor Q21 n is in theon state, the transistor Q21 r is in the off state, the transistor Q29 nis in the off state, and the transistor Q29 r is in the on state. Sincethe node N3 n is charged through the transistor Q21 n and becomes the Hlevel (VDD−Vth), the transistor Q3 n is turned on. In addition, sincethe node N3 r is kept at the L level (VSS) by the transistor Q29 r inthe on state, the transistor Q3 r is kept in the off state.

In the pull-down drive circuit 22, the potential of the drain (firstvoltage signal terminal T1) of the transistor Q26 is VDD, and thepotential of the source (second voltage signal terminal T2) of thetransistor Q27 is VSS.

In this state, the unit shift register SR_(k) in FIG. 21 is equivalentto the circuit in FIG. 19. That is, the transistors Q3 n, Q8 n, and Q21n in the pull-up drive circuit 21 perform the same operations as thoseof the transistors Q3, Q8, and Q21 in FIG. 19, respectively, and thetransistors Q3 r and Q21 r do not contribute to the operation of theunit shift register SR_(k) (the transistors Q3 r and Q21 r are kept inthe off state. Although the transistor Q8 is in the on state while theselection signal G_(k+2) in the second next row is activated, the nodeN4 is at the L level at that time (refer to FIG. 20)). Therefore, thetransistor Q3 n functions as a first charging circuit to charge the nodeN1 in response to the activation of the selection signal G_(k−2) in thesecond previous row. In addition, transistor Q21 n functions as a firstcharging/discharging circuit which charges the node N3 n after theselection signal G_(k−2) in the second previous row is activated andbefore the clock signal CLK3 is activated, and then discharges the nodeN3 n before the clock signal CLK3 is inactivated.

In addition, the pull-down drive circuit 22 operates similar to that inFIG. 19. That is, the pull-down drive circuit 22 puts the node N2 to theL level in response to the activation of the selection signal G_(k−2) inthe second previous row, and puts the node N2 to the H level in responseto the activation of the selection signal G_(k+2) in the second nextrow. In addition, after the node N2 has been put to the H level, thestate is kept by a half latch circuit composed of the transistors Q6, Q7and Q28 until the selection signal G_(k−2) in the second previous row isactivated in the next frame.

Therefore, at the time of forward shift, the unit shift register SR_(k)becomes the set state (the transistor Q1 is in the on state, and thetransistor Q2 is in the off state) in response to the activation of theselection signal G_(k−2) inputted from the second previous row to thefirst input terminal IN1 and the clock signal CLK3, and becomes thereset state (the transistor Q1 is in the off state, and the transistorQ2 is in the on state) in response to the activation of the selectionsignal G_(k+2) inputted from the second next row to the third inputterminal IN3. Therefore, the unit shift register SR_(k) functions as theunit shift register performing the forward shift.

At the time of backward shift, since the first voltage signal Vn is atthe L level (VSS) and the second voltage signal Vr is at the H level(VDD), in the pull-up drive circuit 21, the transistor Q21 n is in theoff state, the transistor Q21 r is in the on state, the transistor Q29 nis in the on state, and the transistor Q29 r is in the off state. Sincethe node N3 r is charged through the transistor Q21 r and becomes the Hlevel (VDD−Vth), the transistor Q3 r is turned on. In addition, sincethe node N3 n is kept at the L level (VSS) by the transistor Q29 n inthe on state, the transistor Q3 n is kept in the off state.

In the pull-down drive circuit 22, the potential of the drain (firstvoltage signal terminal T1) of the transistor Q26 is VSS, and thepotential of the source (second voltage signal terminal T2) of thetransistor Q27 is VDD.

In this state, contrary to the forward shift, the transistors Q3 r, Q8r, and Q21 r in the pull-up drive circuit 21 of the unit shift registerSR_(k) perform the same operations as those of the transistors Q3, Q8,and Q21 in FIG. 19, respectively, and the transistors Q3 n, Q8 n, andQ21 n do not contribute to the operation of the unit shift registerSR_(k). That is, the transistor Q3 r functions as a second chargingcircuit to charge the node N1 in response to the activation of theselection signal G_(k+2) in the second next row. In addition, transistorQ21 r functions as a second charging/discharging circuit which chargesthe node N3 n from when the selection signal G_(k+2) in the second nextrow is activated till when the clock signal CLK2 is activated, and thendischarges the node N3 n before the clock signal CLK2 is inactivated.

In addition, contrary to the forward shift, the pull-down drive circuit22 puts the node N2 to the L level in response to the activation of theselection signal G_(k+2) in the second next row, and puts the node N2 tothe H level in response to the activation of the selection signalG_(k−2) in the second previous row. In addition, after the node N2 hasbeen put to the H level, the state is kept by a half latch circuitcomposed of the transistors Q6, Q7 and Q28 until the selection signalG_(k−2) in the second previous row is activated in the next frame.

Therefore, at the time of backward shift, the unit shift register SRbecomes the set state (the transistor Q1 is in the on state, and thetransistor Q2 is in the off state) in response to the activation of theselection signal G_(k+2) inputted from the second next row to the thirdinput terminal IN3, and becomes the reset state (the transistor Q1 is inthe off state, and the transistor Q2 is in the on state) in response tothe activation of the selection signal G_(k−2) inputted from the secondprevious row to the first input terminal IN1. Therefore, the unit shiftregister SR_(k) functions as the unit shift register performing thebackward shift.

In the unit shift register SR according to this embodiment, theparasitic capacitances of the transistors Q3 n and Q3 r (such as thecapacitance between the gate and the channel and the overlap capacitancebetween the gate and the source/drain) functions as a boosting means forthe nodes N3 n and N3 r. Thus, when the transistor Q3 n or Q3 rprecharges the node N1, its gate potential increases, and thetransistors Q3 n and Q3 r operate in the unsaturated region. Thus, it isnot necessary to separately provide the MOS capacitor element to boostthe nodes N3 n and N3 r. Therefore, the gate line drive circuit small inarea can be implemented. As a result, the gate line drive circuitcapable of bidirectional shifting, which is smaller than those in thesecond and third embodiments, can be implemented.

Eighth Embodiment

As shown in FIG. 2 and FIG. 12, the description above illustrates anexample in which the odd driver 30 a and the even driver 30 b arearranged with the liquid crystal array section 10 sandwichedtherebetween. Accordingly, a formation region of the gate line drivecircuit 30 on the substrate can be efficiently used. However, a largerlength is required for wiring (clock wiring) for supplying the clocksignals CLK1 to CLK3 to the unit shift registers SR in the gate linedrive circuit 30, which increases a parasitic resistance and a parasiticcapacitance of the clock wiring. As a result, if delay is caused in theclock signal, higher speed operation of the gate line drive circuit 30may be hindered.

Therefore, in this embodiment, assuming a case where high speedoperation is emphasized, the unit shift resisters SR in the gate linedrive circuit 30 (odd driver 30 a and even driver 30 b) are all formedon one side of the liquid crystal array section 10, to thereby reducethe length of clock wiring. In particular, the length of the clockwiring can be made shorter by disposing the gate line drive circuit 30on a side closer to the clock signal generator 31 compared with theposition of the liquid crystal array section 10, which is effective.

FIG. 22 and FIG. 23 are block diagrams showing a configuration of a gateline drive circuit according to an eighth embodiment. For example, inthe case where the gate line drive circuit 30 is composed of the unitshift registers SR which perform only forward shift as shown in FIG. 19,a configuration of FIG. 22 is obtained (relationship of connectionbetween respective components is the same as in FIG. 2). Alternatively,in a case where the gate line drive circuit 30 is composed of the unitshift registers SR which perform bidirectional shift as shown in FIG.21, a configuration of FIG. 23 is obtained (relationship of connectionbetween respective components is the same as in FIG. 12).

According to this embodiment, the clock wiring can be reduced, whereby aparasitic resistance and a parasitic capacitance can be shortened, whichprevents delay of a clock signal. Therefore, this embodiment isconducive to higher speed operation of the gate line drive circuit 30.

While the invention has been shown and described in detail, theforegoing description is in all aspects illustrative and notrestrictive. It is therefore understood that numerous modifications andvariations can be devised without departing from the scope of theinvention.

What is claimed is:
 1. A semiconductor device comprising: a MOScapacitor element having two terminals which overlap with one another,wherein a first terminal of the two terminals of said MOS capacitorelement is a gate electrode of an a-Si (amorphous silicon) transistor, asecond terminal of the two terminals of the MOS capacitor element is atleast one current electrode of said a-Si transistor, the gate electrodeof said a-Si transistor is defined by a gate length and a gate width,the gate length of said a-Si transistor is longer than the gate width ofsaid a-Si transistor, and an active layer of said a-Si transistor isdisposed between the two terminals of the MOS capacitor element.
 2. Thesemiconductor device according to claim 1, wherein said a-Si transistorcomprises two current electrodes, the second terminal of said MOScapacitor element is one of said two current electrodes of said a-Sitransistor, and the other of said two current electrodes of said a-Sitransistor in a floating state or is supplied with a constant voltage.3. The semiconductor device according to claim 1, wherein said a-Sitransistor comprises two current electrodes, and the second terminal ofsaid MOS capacitor element is both of said two current electrodes ofsaid a-Si transistor.
 4. A semiconductor device comprising: a MOScapacitor element having two terminals which overlap with one another,wherein a first terminal of the two terminals of said MOS capacitorelement is a gate electrode of an a-Si (amorphous silicon) transistor, asecond terminal of the two terminals of said MOS capacitor element isonly one of two current electrodes of said a-Si transistor, a firstcurrent electrode of the two current electrodes of said a-Si transistoris defined by a first width, a second current electrode of the twocurrent electrodes of said a-Si transistor is defined by a second width,the first width is different from the second width, the second terminalof said MOS capacitor element is only the one of the first currentelectrode and the second current electrode having a width narrower thana width of the other of the first current electrode and the secondcurrent electrode, and an active layer of said a-Si transistor isdisposed between the two terminals of the MOS capacitor element.
 5. Thesemiconductor device according to claim 4, wherein the one of the firstcurrent electrode and the second current electrode having a width longerthan a width of the other of the first current electrode and the secondcurrent electrode is in the floating state or is supplied with theconstant voltage.
 6. The semiconductor device according to claim 1,wherein the semiconductor device is a shift register.
 7. Thesemiconductor device according to claim 4, wherein the semiconductordevice is a shift register.